Frequency modulation and pulse skipping mode voltage controller

ABSTRACT

A voltage converter can be switched among two or more modes to produce an output voltage tracking a reference voltage that can be of an intermediate level between discrete levels corresponding to the modes. One or more voltages generated from a power supply voltage, such as a battery voltage, can be compared with the reference voltage to determine whether to adjust the mode. The reference voltage can be independent of the power supply voltage. Further, the voltage converter may implement frequency modulation and a pulse skipping mode to improve the efficiency of switching operational states of the voltage converter.

RELATED APPLICATIONS

This disclosure claims priority to the following provisionalapplications: U.S. Provisional Application No. 62/057,465, which wasfiled on Sep. 30, 2014 and is titled “VARIABLE SWITCHED DC-TO-DC VOLTAGECONVERTER USING PULSE SKIPPING MODE AND FREQUENCY MODULATION;” U.S.Provisional Application No. 62/057,477, which was filed on Sep. 30, 2014and is titled “FREQUENCY MODULATION BASED VOLTAGE CONTROLLERCONFIGURATION;” and U.S. Provisional Application No. 62/057,627, whichwas filed on Sep. 30, 2014 and is titled “FREQUENCY MODULATION AND PULSESKIPPING MODE VOLTAGE CONTROLLER,” the disclosures of which are eachexpressly incorporated by reference herein in their entirety. Further,this application is related to U.S. application Ser. No. ______, whichwas filed on Sep. 29, 2015 and is titled “VARIABLE SWITCHED DC-TO-DCVOLTAGE CONVERTER USING PULSE SKIPPING MODE AND FREQUENCY MODULATION;”and U.S. application Ser. No. ______, which was filed on Sep. 29, 2015and is titled “FREQUENCY MODULATION BASED VOLTAGE CONTROLLERCONFIGURATION,” the disclosures of which are each expressly incorporatedby reference herein in their entirety.

BACKGROUND

1. Technical Field

The disclosed technology relates to electronic systems and, inparticular, to DC-to-DC voltage converters.

2. Description of Related Technology

One type of device that converts one direct current (“DC”) voltage levelto another DC voltage level may be referred to as a DC-to-DC converter(DC-DC converter). DC-DC converters can be included in battery-operateddevices such as mobile telephones, laptop computers, etc., in which thevarious subsystems of the device require several discrete voltagelevels. In some types of devices, such as a mobile telephone thatoperates in a number of different modes, it can be desirable to supplycertain elements, such as power amplifiers, with a supply voltage at amore efficient level for the mode of operation, rather than waste powerand accordingly drain the battery prematurely. In such devices, it canbe desirable to employ a DC-DC converter that can generate a number ofdiscrete voltage levels.

Some example DC-DC converters include switched-mode DC-DC converters andDC-DC converters that employ pulse-width modulation (PWM). Switched-modeDC-DC converters can convert one DC voltage level to another DC voltagelevel by storing the input energy momentarily or temporarily ininductors and/or capacitors and then releasing that energy to the outputat a different voltage. The switching circuitry can thus continuouslyswitch between two states or phases: a first state in which a network ofinductors and/or capacitors is charging, and a second state in which thenetwork is discharging. The switching circuitry can generate an outputvoltage that is a fixed fraction of the battery voltage, such asone-third, one-half, two-thirds, etc., where a mode selection signal canbe provided as an input to the switching circuitry to control which ofthe fixed fractions is to be employed. Different configurations of thenetwork of inductors and/or capacitors can be selected by manipulatingswitches in the network based on the mode selection signal.

The number of discrete output voltages that a switched-mode DC-DCconverter can generate can be related to the number of inductors and/orcapacitors in the switching circuitry. In a portable, handheld devicesuch as a mobile telephone it can be desirable to minimize size andweight. A DC-DC converter having a relatively large number of inductorsand/or capacitors may not be conducive to minimizing the size and weightof a mobile telephone. A PWM-based DC-DC converter can generate a largernumber of discrete voltages than a switched-mode DC-DC converter withoutemploying significantly more inductors, capacitors and/or other circuitelements. However, a PWM based DC-DC converter can generate a relativelylarge spectrum of spurious output signals that can adversely affect theoperation of a mobile telephone or other frequency-sensitive device.Filters having relatively large capacitances and/or inductances can beincluded in a PWM-based DC-DC converter to minimize these spurioussignals, but large filter capacitors and/or inductors can be undesirablefor at least the reasons relating to size and weight described above.

SUMMARY

Aspects of this disclosure relate to a voltage converter that caninclude a switch matrix including a set of switches configurable into aplurality of states corresponding to a plurality of voltage levelsassociated with a supply voltage. The set of switches may be inelectrical communication with a supply voltage. Further, the switchmatrix can be configured to adjust a state of at least one of the set ofswitches based at least in part on one or more switch control signals.In addition, the switch matrix may be further configured to output afirst voltage. Moreover, the voltage converter can include an oscillatorconfigured to generate an oscillator signal at a frequency and a clockgenerator configured to generate a clock signal based on the oscillatorsignal. In addition, the voltage converter may include control logicconfigured to produce the one or more switch control signals based atleast in part on a comparison between the first voltage and a secondvoltage. The second voltage may be supplied as an input to the voltageconverter. Further, the control logic can be further configured togenerate a frequency control signal for modifying the frequency of theoscillator. The frequency control signal may be based at least in parton the comparison between the first voltage and the second voltage.

In some implementations, the voltage converter further includes afrequency modulation (FM) controller configured to modify the frequencyof the oscillator to generate a modified oscillator signal. The FMcontroller, in some cases, is further configured to supply the modifiedoscillator signal to the clock generator. In addition, the FM controllermay be further configured to modify the frequency of the oscillatorbased at least in part on the comparison between the first voltage andthe second voltage. In some cases, the FM controller is furtherconfigured to reduce the frequency of the oscillator as the differencebetween the first voltage and the second voltage decreases. Moreover,the FM controller can be further configured to reduce the frequency to adefault frequency upon the first voltage exceeding the second voltage.This default frequency may correspond to a threshold frequency foroperation of the oscillator.

Some designs of the FM controller include a voltage controlled currentsource (VCCS) configured to provide a current signal to the oscillator.The VCCS may be an operational transconductance amplifier (OTA).Further, the current signal may be based at least in part on thecomparison between the first voltage and the second voltage.

In some implementations, the voltage converter further includes at leastone capacitor in electrical communication with the switch matrix. The atleast one capacitor can be configured to enable the plurality of voltagelevels. Further, the capacitor may be charged or discharged based atleast in part on a configured state from the plurality of states of theswitch matrix.

Other aspects of this disclosure relate to a wireless device that caninclude a battery configured to power the wireless device and a poweramplifier configured to amplify a radio frequency (RF) input signal andto generate an amplified RF output signal. Further, the wireless devicemay include a voltage converter including a switch matrix, anoscillator, a clock generator, and control logic. The switch may includea set of switches configurable into a plurality of states correspondingto a plurality of voltage levels associated with a supply voltage.Moreover, the set of switches may be in electrical communication with asupply voltage. In addition, the switch matrix can be configured toadjust a state of at least one of the set of switches based at least inpart on one or more switch control signals. Further, the switch matrixmay be further configured to output a first voltage. The oscillator canbe configured to generate an oscillator signal at a frequency and theclock generator can be configured to generate a clock signal based onthe oscillator signal. Moreover, the control logic can be configured toproduce the one or more switch control signals based at least in part ona comparison between the first voltage and a second voltage. The secondvoltage can be supplied as an input to the voltage converter. Inaddition, the control logic can be further configured to generate afrequency control signal for modifying the frequency of the oscillator.The frequency control signal may be based at least in part on thecomparison between the first voltage and the second voltage.

In some implementations, the set of switches are field-effecttransistors (FETs). In addition, the voltage converter may furtherinclude a frequency modulation (FM) controller configured to modify thefrequency of the oscillator to generate a modified oscillator signal andto supply the modified signal to the clock generator. The FM controllermay include a voltage controlled current source (VCCS) configured toprovide a current signal to the oscillator, the modified signal suppliedto the clock generator output by the oscillator and based at least inpart on the current signal. Moreover, the voltage converter may beconfigured to transition into a pulse skipping mode based at least inpart on the comparison of the first voltage with the second voltage.

Yet other aspects of the present disclosure relate to a multi-chipmodule that can include a power amplifier die including one or morepower amplifiers and a controller die including a power amplifier biascontroller and a voltage converter. The voltage controller can include aswitch matrix, an oscillator, a clock generator, and control logic. Theswitch may include a set of switches configurable into a plurality ofstates corresponding to a plurality of voltage levels associated with asupply voltage. The set of switches may be in electrical communicationwith a supply voltage. Further, the switch matrix may be configured toadjust a state of at least one of the set of switches based at least inpart on one or more switch control signals. The switch matrix can befurther configured to output a first voltage, the oscillator may beconfigured to generate an oscillator signal at a frequency, and theclock generator may be configured to generate a clock signal based onthe oscillator signal. Further, the control logic can be configured toproduce the one or more switch control signals based at least in part ona comparison between the first voltage and a second voltage. The secondvoltage may be supplied as an input to the voltage converter. Inaddition, the control logic may be further configured to generate afrequency control signal for modifying the frequency of the oscillator.The frequency control signal may be based at least in part on thecomparison between the first voltage and the second voltage.

In some implementations, the voltage converter further includes afrequency modulation (FM) controller configured to modify the frequencyof the oscillator to generate a modified oscillator signal and to supplythe modified signal to the clock generator. Further, the FM controllermay include a voltage controlled current source (VCCS) configured toprovide a current signal to the oscillator. The modified signal can besupplied to the clock generator output by the oscillator and may bebased at least in part on the current signal. Moreover, the voltageconverter may be configured to transition into a pulse skipping modebased at least in part on the comparison of the first voltage with thesecond voltage.

One aspect of this disclosure relates to an apparatus that can include aswitch matrix configured to output a plurality of voltage levels. Theapparatus can further include an oscillator configured to generate anoscillator signal at a frequency and a clock generator configured togenerate a clock signal based on the oscillator signal. In addition, theapparatus can include control logic that includes a first input coupledto an output voltage and a second input coupled to a reference voltage.The output voltage may correspond to a feedback voltage output by theswitch matrix. The control logic can further include a comparatorconfigured to compare the output voltage with the reference voltage.Moreover, the control logic can be configured to, based at least in parton the comparison, modify the frequency of the oscillator signal andgenerate the one or more mode control signals for setting a mode of theswitch matrix.

In some cases, the control logic may further include a voltagecontrolled current source (VCCS) configured to modify the frequency ofthe oscillator signal. The VCCS may be an operational transconductanceamplifier (OTA).

Moreover, the control logic may be further configured to modify thefrequency of the oscillator signal based at least in part on adifference between the output voltage and the reference voltage when thereference voltage exceeds the output voltage. In addition, the controllogic may be further configured to reduce the frequency of theoscillator signal as the difference between the output voltage and thereference voltage is reduced. Further, the control logic may be furtherconfigured to set the oscillator signal to a minimum threshold frequencywhen the output voltage exceeds the reference voltage. In addition, thecontrol logic may be further configured to deactivate the switch matrixwhen the output voltage exceeds the reference voltage.

In certain implementations, the apparatus also includes a plurality ofcapacitive circuit elements operatively coupled to the switch matrix. Insome such implementations, the switch matrix is further configured toimplement a plurality of modes, each mode having a first phaseconfiguration in which at least one of the plurality of capacitivecircuit elements is charged and a second phase configuration in whichthe at least one of the plurality of capacitive circuit elements isdischarged. Further, the control logic may cause causes the switchmatrix to implement a mode from the plurality of modes having the secondphase configuration when the output voltage exceeds the referencevoltage.

Moreover, the comparator may be a hysteresis comparator. In addition,the reference voltage may correspond to a target voltage for the outputvoltage. This reference voltage may be specified based on an operatingenvironment for the apparatus.

Another aspect of this disclosure relates to a wireless device that caninclude a battery configured to power the wireless device and a poweramplifier configured to amplify a radio frequency (RF) input signal andto generate an amplified RF output signal. Further, the wireless devicecan include a direct current to direct current (DC-DC) voltage converterconfigured to generate an output voltage to control the power amplifierso as to improve power efficiency. The DC-DC voltage converter mayinclude a switch matrix, an oscillator, a clock generator, and controllogic. The switch matrix may be configured to output a plurality ofvoltage levels. The oscillator can be configured to generate anoscillator signal at a frequency and the clock generator can beconfigured to generate a clock signal based on the oscillator signal.Further, the control logic may include a first input coupled to anoutput voltage and a second input coupled to a reference voltage. Theoutput voltage may correspond to a feedback voltage output by the switchmatrix. Moreover, the control logic may further include a comparatorconfigured to compare the output voltage with the reference voltage. Thecontrol logic may be configured to, based at least in part on thecomparison, modify the frequency of the oscillator signal and generatethe one or more mode control signals for setting a mode of the switchmatrix.

The control logic, in some designs, further includes a voltagecontrolled current source (VCCS) configured to modify the frequency ofthe oscillator signal. Further, the control logic may be furtherconfigured to modify the frequency of the oscillator signal by reducingthe frequency of the oscillator signal to a minimum operating frequencywhen the output voltage exceeds the reference voltage and to deactivatethe switch matrix when the output voltage exceeds the reference voltage.Moreover, the control logic can be further configured to modify thefrequency of the oscillator signal by reducing the frequency of theoscillator signal in proportion to a difference between the outputvoltage and the reference voltage when the reference voltage exceeds theoutput voltage.

In some implementations, the DC-DC voltage converter is configured totransition into a pulse skipping mode based at least in part on thecomparison of the output voltage with the reference voltage. Further, atleast one of the one or more switches may have a plurality ofsub-switches. At least one of the plurality of sub switches may beconfigured to change states based at least in part on an amount ofcurrent provided by the DC-DC voltage converter. Further, in somedesigns the comparator of the control logic is a hysteresis comparator.

A further aspect of this disclosure relates to a method that includesgenerating a first voltage using a DC-DC voltage converter. The methodmay further include comparing the first voltage generated by the DC-DCvoltage converter with a second voltage that is independent of the firstvoltage to obtain a comparison signal. In addition, the method mayinclude setting a frequency of an oscillator signal based at leastpartially on the comparison signal. The oscillator signal may be used togenerate a clock signal for the DC-DC voltage converter.

The method may further include deactivating a switch matrix of the DC-DCvoltage converter responsive to the comparison signal. In addition, themethod may include setting the second voltage based on an operatingenvironment of a device including the DC-DC voltage converter. In somecases, the comparing of the first voltage with the second voltage isfurther based on a hysteresis value.

Aspects of this disclosure relate to an apparatus including a switchmatrix that includes switches configurable into a plurality of statescorresponding to a plurality of voltage levels of an output voltage. Theswitch matrix may be configured to adjust a state of at least one of theswitches based at least in part on one or more switch control signals.Further, the apparatus may include control logic configured to receive afirst voltage and a second voltage, compare the first voltage with thesecond voltage, modify a frequency of an oscillator signal based atleast in part on the result of the comparison, and generate a switchcontrol signal from the one or more switch control signals.

Further, the control logic may be further configured to reduce thefrequency of the oscillator signal as a difference between the firstvoltage and the second voltage decreases. In addition, the control logicmay be further configured to reduce the frequency of the oscillatorsignal to a minimum frequency when the difference between the firstvoltage and the second voltage is negative. Moreover, the control logicmay be further configured to transition the apparatus into a pulseskipping mode based at least in part on the comparison of the firstvoltage with the second voltage.

In certain implementations, the apparatus includes an oscillatorconfigured to generate the oscillator signal in response to a currentgenerated based at least in part on the result of comparison of thefirst voltage and the second voltage. In addition, the apparatus mayalso include a voltage controlled current source (VCCS) configured togenerate the current and provide the current to the oscillator. The VCCSmay be an operational transconductance amplifier (OTA).

Some designs of the apparatus may include a clock generator configuredto generate one or more clock signals for the switch matrix. The one ormore clock signals may be based at least in part on the oscillatorsignal. Further, the apparatus may include a switch controllerconfigured to configure the switch matrix based on the switch controlsignal. In some such cases, the control logic is further configured toprovide the switch control signal to the switch controller. Moreover,the apparatus may also include a capacitor in electrical communicationwith the switch matrix. The switch controller may be further configuredto charge or discharge the capacitor based on the switch control signal.

Other aspects of this disclosure relate to a wireless device that caninclude a battery configured to power the wireless device and a poweramplifier configured to amplify a radio frequency (RF) input signal andto generate an amplified RF output signal. Further, the wireless devicecan include a voltage controller including a switch matrix and controllogic. The switch matrix may include switches configurable into aplurality of states corresponding to a plurality of voltage levels of anoutput voltage. Moreover, the switch matrix may be configured to adjusta state of at least one of the switches based at least in part on one ormore switch control signals. The control logic can be configured toreceive a first voltage and a second voltage, compare the first voltagewith the second voltage, modify a frequency of an oscillator signalbased at least in part on the result of the comparison, and generate aswitch control signal from the one or more switch control signals.

The control logic, in certain cases, may be further configured to reducethe frequency of the oscillator signal as a difference between the firstvoltage and the second voltage decreases. Further, the control logic maybe further configured to reduce the frequency of the oscillator signalto a minimum frequency when the difference between the first voltage andthe second voltage is negative. In addition, the control logic may befurther configured to transition the apparatus into a pulse skippingmode based at least in part on the comparison of the first voltage withthe second voltage.

In some implementations, the voltage controller further includes anoscillator and a voltage controlled current source (VCCS). This VCCS maybe configured to generate a current based at least in part on the resultof the comparison and to provide the current to the oscillator. Further,the oscillator may be configured to generate the oscillator signal basedat least in part on the current. In some cases, the VCCS is anoperational transconductance amplifier (OTA).

Yet other aspects of this disclosure relate to a multi-chip module thatcan include a power amplifier die including one or more poweramplifiers. Further, the multi-chip module may include a controller dieincluding a power amplifier bias controller and a voltage converter. Thevoltage converter may include a switch matrix and control logic. Theswitch matrix can include switches configurable into a plurality ofstates corresponding to a plurality of voltage levels of an outputvoltage. Moreover, the switch matrix may be configured to adjust a stateof at least one of the switches based at least in part on one or moreswitch control signals. The control logic can be configured to receive afirst voltage and a second voltage, compare the first voltage with thesecond voltage, modify a frequency of an oscillator signal based atleast in part on the result of the comparison, and generate a switchcontrol signal from the one or more switch control signals.

In some designs, the control logic of the voltage converter may befurther configured to set the frequency of the oscillator signal basedat least in part on a difference between the first voltage and thesecond voltage. Moreover, the control logic may be further configured totransition the apparatus into a pulse skipping mode based at least inpart on the comparison of the first voltage with the second voltage. Inaddition, in some designs, the voltage controller further includes anoscillator and a voltage controlled current source (VCCS). The VCCS maybe configured to generate a current based at least in part on the resultof the comparison and to provide the current to the oscillator. Thisoscillator can be configured to generate the oscillator signal based atleast in part on the current provided by the VCCS.

BRIEF DESCRIPTION OF THE DRAWINGS

Throughout the drawings, reference numbers are re-used to indicatecorrespondence between referenced elements. The drawings are provided toillustrate embodiments of the inventive subject matter described hereinand not to limit the scope thereof.

FIG. 1A is a schematic block diagram of an example voltage converterthat can implement frequency modulation (FM).

FIG. 1B is a schematic block diagram of an illustrative wireless device.

FIG. 2 is a schematic block diagram of an illustrative multi-chipmodule.

FIGS. 3A and 3B are schematic block diagrams of voltage convertersaccording to certain embodiments.

FIG. 4A is a circuit diagram illustrating an example switch matrix ofFIG. 3A and FIG. 3B in a first phase configuration of a first modeconfiguration.

FIG. 4B is a circuit diagram illustrating the example switch matrix ofFIG. 3A and FIG. 3B in a second phase configuration of the first modeconfiguration.

FIG. 5A is a circuit diagram illustrating the example switch matrix ofFIG. 3A and FIG. 3B in a first phase configuration of a second modeconfiguration.

FIG. 5B is a circuit diagram illustrating the example switch matrix ofFIG. 3A and FIG. 3B in a second phase configuration of the second modeconfiguration.

FIG. 6A is a circuit diagram illustrating the example switch matrix ofFIG. 3A and FIG. 3B in a first phase configuration of a variant of thesecond mode configuration.

FIG. 6B is a circuit diagram illustrating the example switch matrix ofFIG. 3A and FIG. 3B in a second phase configuration of the variant ofthe second mode configuration.

FIG. 7A is a circuit diagram illustrating the example switch matrix ofFIG. 3A and FIG. 3B in a first phase configuration of a third modeconfiguration.

FIG. 7B is a circuit diagram illustrating the example switch matrix ofFIG. 3A and FIG. 3B in a second phase configuration of the third modeconfiguration.

FIG. 8A is a circuit diagram of an example comparator circuit of FIG. 3Aand FIG. 3B.

FIG. 8B shows another example switch matrix in combination withcapacitive circuit elements.

FIG. 8C illustrates an example schematic of a bias control circuit for apower amplifier.

FIG. 9A is a truth table associated with example mode selection logic ofFIG. 3A and FIG. 3B.

FIG. 9B is an example signal graph illustrating the application of theexample mode selection logic reflected in the truth table of FIG. 9A.

FIG. 10 is a circuit diagram illustrating example switch control logicshown in FIG. 3A and FIG. 3B.

FIG. 11 is a timing diagram illustrating an example of operation of thevoltage converter of FIG. 3A and FIG. 3B.

FIG. 12 is a flow diagram illustrating an example method of operation ofthe voltage converter of FIG. 3A and FIG. 3B.

FIG. 13 is a schematic block diagram of a voltage converter that canoperate using both PSM and FM in accordance with certain embodiments.

FIG. 14A is another schematic block diagram of a voltage converter thatcan operate using both PSM and FM according to certain embodiments.

FIG. 14B is a schematic block diagram illustrating additional details ofthe voltage converter presented in FIG. 14A.

FIG. 15 is a graph comparing the difference in driver circuit currentdrawn by a voltage converter operating with PSM and a voltage converteroperating with PSM and FM.

FIG. 16 is a graph comparing the efficiency of a DC-DC voltage converteroperating with PSM with a DC-DC voltage converter operating with PSM andFM.

DETAILED DESCRIPTION

Generally described, aspects of the present disclosure relate to DC-DCvoltage conversion. More specifically, some implementations relate tovariable switched capacitor DC-DC converters. Using the voltageconversion systems, apparatus, and methods described herein, electronicsystems, such as power amplifier systems, can operate more efficientlyand/or consume less power. For instance, power amplifier biasing can bemade more efficient. A switched capacitor architecture can utilizefeedback to dynamically select switch mode and generate a controlvoltage. Switch sizes and/or switching frequency can be dynamicallyadjusted.

DC-DC converters that use pulse width modulation (PWM) architectures cangenerate undesirable output spectrum including noise. In such PWMarchitectures, the output noise spectrum may require an LC filter thatincludes a large inductive element that is expensive to implement andcan radiate energy. Switched capacitor architectures can remove the needfor LC filters that include large inductive elements. However, someswitched capacitor architectures may be configured to generate discretevoltage levels that are a function of an input supply voltage and acapacitor division ratio.

Advantageously, a switched DC-DC capacitor architecture can use feedbackto dynamically select a division ratio of a voltage converter bytoggling between various output states. As a result, an intermediatevoltage can be generated over a continuous range between a lowestdivision mode and a highest division mode.

The methods, systems, and apparatus for voltage conversion describedherein may be able to achieve one or more of the following advantageousfeatures, among others. An output noise spectrum can be reduced comparedto conventional PWM architectures. The switched capacitor architecturecan be modeled as an RC filter with a corner frequency related to aswitching frequency and capacitors. Low pass characteristics of thefilter can allow for elimination of the conventionally large inductorsused in an LC filter network and/or support integration into a smallpackage foot print or co-integration with power amplifier products.

The use of feedback described herein can further reduce the noisespectrum on an output of a voltage converter. The feedback can alsoimprove the tolerance of a voltage converter to noise and/or jitter onthe output of the voltage regulator.

The switch control methods described herein can use comparator circuitsto optimize effective switch size based on reference voltage and/orpower amplifier output power levels. Alternatively or additionally, thecomparator circuits can be used to optimize switching frequency based onreference voltage and/or power amplifier output power levels.

One or more outputs of the comparator circuit can be used to optimize abias current based on different reference voltage. The bias profile canalso track the reference voltage like a power amplifier collectorvoltage.

The feedback methods and methods to adjust switch size, switchingfrequency, and power amplifier biasing according to an input referencevoltage can make the power amplifier more efficient. This can lead toreduced power consumption and extended battery life in systems poweredby a battery.

The headings provided herein are for convenience only and do notnecessarily affect the scope or meaning of the claimed invention.

Overview of Voltage Converter

FIG. 1A is a schematic block diagram of an example voltage converter 100that can implement frequency modulation (FM). As will be described inmore detail herein, the voltage converter 100 can implement both a pulseskipping mode (PSM) and frequency modulation (FM). Advantageously,implementing both PSM and FM enables the voltage converter 100 to moreefficiently achieve a desired output voltage compared to a voltageconverter that does not implement both PSM and FM. Further, the voltageconverter 100 is able to avoid the problems or drawbacks associated withdeactivating the oscillator of the voltage converter, such as the timerequired to restart or reinitialize the oscillator when an outputvoltage drops below a desired value.

As is described in more detail below, the voltage converter 100 cangradually adjust the frequency of the clock signal by basing, at leastin part, the oscillator output on a current generated by an operationaltrans-conductance amplifier (OTA). The OTA can base, at least in part,its generated current on a difference between the output voltage and thedesigned voltage (e.g., a reference voltage). As the difference narrowsor widens, the current output by the OTA can change. Further, detailsregarding the implementation and advantages of the voltage converterusing both PSM and FM are described in more detail below.

Overview of Wireless Devices

Any of the methods, apparatus, and computer-readable media for DC-DCvoltage conversion described herein can be implemented in a variety ofelectronic devices, such as a wireless device, which can also bereferred to as a mobile device. FIG. 1B schematically depicts a wirelessdevice 1. Examples of the wireless device 1 include, but are not limitedto, a cellular phone (e.g., a smart phone), a laptop, a tablet computer,a personal digital assistant (PDA), a wearable device (e.g., asmartwatch or an optical head-mounted device), an electronic bookreader, and a portable digital media player. For instance, the wirelessdevice 1 can be a multi-band and/or multi-mode device such as amulti-band/multi-mode mobile phone configured to communicate using, forexample, a Global System for Mobile Communications (GSM) network, a codedivision multiple access (CDMA) network, a 3G network, a 4G network,and/or a long term evolution (LTE) network.

In certain embodiments, the wireless device 1 can include one or more ofa RF front end 2, a transceiver component 3, an antenna 4, one or morepower amplifiers 5, a control component 6, a computer readable medium 7,a processor 8, a battery 9, and a supply control block 10.

The transceiver component 3 can generate RF signals for transmission viathe antenna 4. Furthermore, the transceiver component 3 can receiveincoming RF signals from the antenna 4.

It will be understood that various functionalities associated with thetransmission and receiving of RF signals can be achieved by one or morecomponents that are collectively represented in FIG. 1B as thetransceiver 3. For example, a single component can be configured toprovide both transmitting and receiving functionalities. In anotherexample, transmitting and receiving functionalities can be provided byseparate components.

Similarly, it will be understood that various antenna functionalitiesassociated with the transmission and receiving of RF signals can beachieved by one or more components that are collectively represented inFIG. 1B as the antenna 4. For example, a single antenna can beconfigured to provide both transmitting and receiving functionalities.In another example, transmitting and receiving functionalities can beprovided by separate antennas. In yet another example, different bandsassociated with the wireless device 1 can be provided with differentantennas.

In FIG. 1B, one or more output signals from the transceiver 3 aredepicted as being provided to the antenna 4 via one or more transmissionpaths. In the example shown, different transmission paths can representoutput paths associated with different bands and/or different poweroutputs. For instance, the two example power amplifiers 5 shown canrepresent amplifications associated with different power outputconfigurations (e.g., low power output and high power output), and/oramplifications associated with different bands.

In FIG. 1B, one or more detected signals from the antenna 4 are depictedas being provided to the transceiver 3 via one or more receiving paths.In the example shown, different receiving paths can represent pathsassociated with different bands. For example, the four example pathsshown can represent a quad-band capability supported by some wirelessdevices.

To facilitate switching between receive and transmit paths, the RF frontend 2 can be configured to electrically connect the antenna 4 to aselected transmit or receive path. Thus, the RF front end 2 can providea number of switching functionalities associated with an operation ofthe wireless device 1. In certain embodiments, the RF front end 2 caninclude a number of switches configured to provide functionalitiesassociated with, for example, switching between different bands,switching between different power modes, switching between transmissionand receiving modes, or some combination thereof. The RF front end 2 canalso be configured to provide additional functionality, includingfiltering of signals. For example, the RF front end can include one ormore duplexers.

The wireless device 1 can include one or more power amplifiers 5. RFpower amplifiers can be used to boost the power of a RF signal having arelatively low power. Thereafter, the boosted RF signal can be used fora variety of purposes, included driving the antenna of a transmitter.Power amplifiers 5 can be included in electronic devices, such as mobilephones, to amplify a RF signal for transmission. For example, in mobilephones having an architecture for communicating using the 3G and/or 4Gcommunications standards, a power amplifier can be used to amplify a RFsignal. It can be desirable to manage the amplification of the RFsignal, as a desired transmit power level can depend on how far the useris away from a base station and/or the mobile environment. Poweramplifiers can also be employed to aid in regulating the power level ofthe RF signal over time, so as to prevent signal interference fromtransmission during an assigned receive time slot. A power amplifiermodule can include one or more power amplifiers.

FIG. 1B shows that in certain embodiments, a control component 6 can beprovided, and such a component can be configured to provide variouscontrol functionalities associated with operations of the RF front end2, the power amplifiers 5, the supply control 10, and/or other operatingcomponent(s). Non-limiting examples of the supply control 10 aredescribed herein in greater detail.

In certain embodiments, a processor 8 can be configured to facilitateimplementation of various processes described herein. For the purpose ofdescription, embodiments of the present disclosure may also be describedwith reference to flowchart illustrations and/or block diagrams ofmethods, apparatus (systems) and computer program products. It will beunderstood that each block of the flowchart illustrations and/or blockdiagrams, and combinations of blocks in the flowchart illustrationsand/or block diagrams, may be implemented by computer programinstructions. These computer program instructions may be provided to aprocessor of a general purpose computer, special purpose computer, orother programmable data processing apparatus to produce a machine, suchthat the instructions, which execute via the processor of the computeror other programmable data processing apparatus, create means forimplementing the acts specified in the flowchart and/or block diagramblock or blocks.

In certain embodiments, these computer program instructions may also bestored in a computer-readable memory 7 that can direct a computer orother programmable data processing apparatus to operate in a particularmanner, such that the instructions stored in the computer-readablememory produce an article of manufacture including instructions whichimplement the acts specified in the flowchart and/or block diagram blockor blocks. The computer program instructions may also be loaded onto acomputer or other programmable data processing apparatus to cause aseries of operations to be performed on the computer or otherprogrammable apparatus to produce a computer implemented process suchthat the instructions that execute on the computer or other programmableapparatus provide operations for implementing the acts specified in theflowchart and/or block diagram block or blocks.

The illustrated wireless device 1 also includes the supply control block10, which can be used to provide a power supply to one or more of thepower amplifiers 5. For example, the supply control block 10 can be aDC-to-DC converter. However, in certain embodiments the supply controlblock 10 can include other blocks, such as, for example, an envelopetracker configured to vary the supply voltage provided to the poweramplifiers 5 based upon an envelope of the RF signal to be amplified.

The supply control block 10 can be electrically connected to the battery9, and the supply control block 10 can be configured to vary the voltageprovided to the power amplifiers 5 based on an output voltage of a DC-DCconverter. The battery 9 can be any suitable battery for use in thewireless device 1, including, for example, a lithium-ion battery. Byvarying the voltage provided to the power amplifiers, the powerconsumption of the battery 9 can be reduced, thereby improvingperformance of the wireless device 1.

Overview of Multi-Chip Module

FIG. 2 is a schematic block diagram of a power amplifier module. Theillustrated power amplifier module is a multi-chip module (MCM) 200 thatcan include one or more of a controller die 202, a power amplifier die204, inductor(s) 206, capacitor(s) 208, and an impedance matchingcomponent 210. In some embodiments, the MCM 200 can include one or moreof the PAs 5 and elements of the RF front end 2 from the wireless device1. The multi-chip module 200 can include a plurality of dies and/orother components mounted on a carrier substrate of the module. In someimplementations, the substrate can be a multi-layer substrate configuredto support the dies and/or components and to provide electricalconnectivity to external circuitry when the module is mounted on acircuit board, such as a phone board.

The power amplifier die 204 can receive a RF signal on an input pinRF_(IN) (or RF_IN) of the multi-chip module. The power amplifier die 204can include one or more power amplifiers 5, including, for example,multi-stage power amplifiers configured to amplify the RF signal. Theamplified RF signal can be provided to an output pin RF_(OUT) (orRF_OUT) of the power amplifier module. The impedance matching component210 can be provided on the power amplifier module to aid in reducingsignal reflections and/or other signal distortions. The power amplifierdie 204 can be any suitable die. In some implementations, the poweramplifier die 204 is a gallium arsenide (GaAs) die. In some of theseimplementations, the GaAs die has transistors formed using aheterojunction bipolar transistor (HBT) process.

The multi-chip module 200 can also include a V_(CC) pin, which can beelectrically connected to a controller. The multi-chip module 200 caninclude the inductor(s) 206, which can be formed, for example, by traceson the multi-chip module 200. These traces may be formed using aconductive material applied to the circuit board or printed circuitboard (PCB). The inductor(s) 206 can operate as a choke inductor, andcan be disposed between the supply voltage and the power amplifier die204. In some implementations, the inductor(s) 206 are surface mounted.Additionally, capacitor(s) 208 can be electrically connected in parallelwith the inductor(s) 208 and can be configured to resonate at afrequency near the frequency of a signal received on the pin RF_(IN). Insome implementations, the capacitor(s) 208 include a surface mountedcapacitor.

In some implementations, the controller die 202 can be manufactured on asilicon wafer. In some of these implementations, the controller die 202can be manufactured using complementary metal oxide semiconductor (CMOS)process technology. The controller die 202 can include a power amplifierbias control block 212 and/or a DC-DC converter block 214. The poweramplifier bias control block 212 can be used, for example, to providebias signals to the power amplifier die 204. For example, in a bipolartransistor power amplifier configuration, the controller die 202 can beused to provide a reference voltage for biasing a current mirror used togenerate a base current for the power amplifiers 5, such as a basecurrent for a bipolar transistor. The controller die 202 can also beused to enable and/or disable a power amplifier 5 disposed on the poweramplifier die 204, which can aid in selectively activating a poweramplifier 5 associated with a particular transmission path. For example,the controller die 202 can receive a control signal on a pin CONTROL,and can use the control signal to vary the bias signal provided to thepower amplifier die 204 so as to selectively enable or disable the poweramplifier 5. The controller die 202 can also include a DC-DC converter214, as will be described in more detail herein.

The multi-chip module 200 can be modified to include more or fewercomponents, including, for example, additional power amplifier dies,capacitors and/or inductors. For instance, the multi-chip module 200 caninclude an additional power amplifier die, as well as an additionalcapacitor and inductor configured to operate as a parallel LC circuitdisposed between the additional power amplifier die and the V_(CC) pinof the module. The multi-chip module 200 can be configured to haveadditional pins, such as in implementations in which a separate powersupply is provided to an input stage disposed on the power amplifier die204 and/or implementations in which the multi-chip module 200 operatesover a plurality of bands.

DC-DC Voltage Conversion

As illustrated in FIG. 3A, in an illustrative embodiment, a DC-DCvoltage converter 214 can include two capacitors 12 and 14, a switchmatrix 16, and control logic 17. The control logic 17 can include anysuitable circuit elements configured to adjust states of the switchmatrix 16. The control logic 17 can include a comparator circuit 18 andswitching logic 20. A reference voltage signal V_REF can be provided tothe DC-DC voltage converter 214 as a control input. In certainembodiments, the reference voltage signal V_REF can be generated by adigital-to-analog converter (DAC) that is external to the DC-DCconverter 214. Such a DAC can control an output of the DC-DC converter214 for continuous power amplifier power adjustment. According to someembodiments, the reference voltage V_REF can be bypassed by a relativelylarge capacitor (for example, having a capacitance on the order of 1000pF) to analog ground. Such a bypass capacitor can have a first endelectrically coupled to the reference voltage V_REF provided to an inputto the DC-DC converter 214, and the bypass capacitor can have a secondend coupled to analog ground. As a result, the reference voltage V_REFcan be a relatively noise-free signal. The DC-DC voltage converter 214can produce an output voltage signal V_OUT that can correspond to and/ortrack the reference voltage signal V_REF. In some implementations, theoutput voltage signal V_OUT can be provided to a collector of a bipolartransistor in a power amplifier 5. The DC-DC voltage converter 214 canfurther include a clock signal generator circuit 22 and associatedoscillator 24 that can be activated by an enable signal, ENABLE. Theenable signal can remain active during the operation of the DC-DCvoltage converter 214.

The switch matrix 16 is configurable into one of several modeconfigurations, for example, as described below, in which the capacitors12 and 14 are interconnected in different configurations. Each modeconfiguration can be referred to as a state of the switch matrix 16. Ineach mode configuration, switch matrix 16 can assume either a firstphase configuration, in which the capacitor circuit including theinterconnected capacitors 12 and 14 is charging, or a second phaseconfiguration, in which the capacitor circuit including theinterconnected capacitors 12 and 14 is discharging. The switch matrix 16can provide the output of the capacitor circuit at an output node 26. Inoperation, the switch matrix 16 can alternately switch between the firstand second phase configurations in response to the clock signalgenerated by the clock signal generator 22. Filter circuitry, such as acapacitor 28, can be connected to output node 26 to filter the outputvoltage signal V_OUT.

As described in further detail below, comparator circuit 18 can comparethe reference voltage signal V_REF with one or more other referencesignals generated by a power supply that can be independent of thereference voltage signal V_REF. In response to one or more of thesecomparisons, the comparator circuit 18 can generate a number ofcomparison signals 30. The switching logic 20 can include mode selectionlogic 32 and switch control logic 34. The mode selection logic 32 canreceive the comparison signals 30 and, in response, generate one or moremode selection signals 36. The switch control logic 34 can receive modeselection signals 36 and, in response, generate one or more switchcontrol signals 38.

FIG. 3B shows another illustrative DC-DC converter 214. The DC-DCconverter of FIG. 3B is substantially the same as the DC-DC converter ofFIG. 3A, except that the DC-DC converter of FIG. 3B receives additionalinput signal(s) that can be used to program the DC-DC converter of FIG.3B to operate in either step mode(s) or continuous mode(s). Thus, theDC-DC converter of FIG. 3B can be programmable between step modes andcontinuous mode. For example, providing one or more control signals,such as V_CTRL1 and/or V_CTRL2, to the control logic 17, can generate anoutput voltage V_OUT that can either be a continuous output voltage orone of a plurality of discrete output voltages. More specifically, withtwo digital bits V_CTRL1, V_CTRL2 and a reference voltage V_REF, theDC-DC converter 214 can either produce the continuous output or use twocontrol bits control V_CTRL1, V_CTRL2 to generate 4 discrete outputvoltages. The control bits V_CTRL1, V_CTRL2 can be baseband inputs. Thecontrol bits can drive an analog signal, or two or more digital signals.The control bits can, for example, clock gate circuitry related to thereference voltage V_REF. In some implementations, the DC-DC converter214 may be able to generate both continuous and discrete voltagesconcurrently.

As illustrated in FIGS. 4A, 4B, 5A, 5B, 6A, 6B, 7A and 7B, the switchmatrix 16 can interconnect capacitors 12 and 14 in several differentconfigurations between a voltage potential (e.g., either a supplyvoltage, such as the battery voltage, or ground) and the output node 26.The illustrative switch matrix 16 includes nine switches 40, 42, 44, 46,48, 50, 52, 54 and 56, which are controlled by the switch controlsignals 38 (S1-S9). Although switches 40-56 are shown schematically inFIGS. 4A-7B in the form of controllable, single-pole, single-throw(SPST) switches, they can include any suitable switching devices, suchas field-effect transistors (FETs). For example, in someimplementations, each of switches 40 and 50 can include a P-type FET(PFET), each of switches 46 and 56 can include an N-type FET (NFET), andeach of switches 42, 44, 48, 52 and 54 can include a parallelcombination of a PFET and an NFET. The control terminal (e.g., gate) ofeach FET can receive one of the switch control signals 38 (S1-S9).

Although the switch matrix 16 includes nine switches in the exampleembodiment, which can be arranged as shown, in other embodiments, aswitch matrix can include any suitable number of switches arranged inany other suitable manner. Similarly, although the illustrativeembodiment includes two capacitors 12 and 14, which the switch matrix 16can interconnect as described below, other embodiments can include morethan two capacitors, and a switch matrix can interconnect the capacitorsin a variety of suitable configurations. Furthermore, applying theprinciples and advantages described herein, any suitable number of modescan be implemented using a switch matrix. In addition, although abattery voltage is described, another power supply voltage can be incombination with a battery or as an alternative.

As illustrated in FIGS. 4A-B, in a first configuration, the switchmatrix 16 can interconnect the capacitors 12 and 14 in either the firstphase configuration shown in FIG. 4A or the second phase configurationshown in FIG. 4B. This first configuration can be referred to herein asthe “1/3 mode” because operation in this mode is intended to result inan output voltage signal V_OUT at output node 26 having a voltage levelthat is nominally or on average about one-third of the battery voltageV_BATT, according to some implementations.

As shown in FIG. 4A, in the first phase configuration of the 1/3 mode,the switches 40, 44, 48, 50 and 54 are open, and the switches 42, 46, 52and 56 are closed. The combination of the closed states of switches 42and 46 couples, or electrically connects, capacitor 12 between a groundvoltage potential (e.g., 0 volts) and the output node 26. Thecombination of the closed states of switches 52 and 56 similarlycouples, or electrically connects, capacitor 14 between the groundpotential and output node 26 (i.e., in parallel with capacitor 12).Thus, in the first phase configuration of the 1/3 mode, the capacitorcircuit including capacitors 12 and 14 in parallel with each otherdischarges with respect to output node 26.

As shown in FIG. 4B, in the second phase configuration of the 1/3 mode,the switches 42, 44, 46, 50, 52 and 56 are open, and the switches 40, 48and 54 are closed. The combination of the closed states of switches 40,48 and 54 couples capacitors 12 and 14 in series between a positivevoltage potential, such as a base reference voltage provided by abattery V_BATT or other voltage supply, and output node 26. Thus, in thesecond phase configuration of the 1/3 mode, the capacitor circuitincluding capacitors 12 and 14 in series with each other charges withrespect to output node 26.

As illustrated in FIGS. 5A-B, in a second configuration, the switchmatrix 16 can interconnect capacitors 12 and 14 in either the firstphase configuration shown in FIG. 5A or the second phase configurationshown in FIG. 5B. This second configuration can be referred to herein asthe “1/2A mode” because operation in this mode is intended to result inan output voltage signal V_OUT at output node 26 having a voltage levelthat is nominally or on average about one-half of the battery voltageV_BATT, according to some implementations. Also, as described below,there is a variant of the 1/2A mode, referred to as the 1/2B mode.

As shown in FIG. 5A, in the first phase configuration of the 1/2A mode,the switches 40, 44, 48, 50 and 54 are open, and the switches 42, 46, 52and 56 are closed. The combination of the closed states of switches 42and 46 couples capacitor 12 between ground and output node 26. Thecombination of the closed states of switches 52 and 56 similarly couplescapacitor 14 between ground and output node 26 (i.e., in parallel withcapacitor 12). Thus, in the first phase configuration of the 1/2A mode,the capacitor circuit including capacitors 12 and 14 in paralleldischarges with respect to output node 26.

As shown in FIG. 5B, in the second phase configuration of the 1/2A mode,the switches 42, 46, 48, 52 and 56 are open, and the switches 40, 44, 50and 54 are closed. The combination of the closed states of switches 40and 44 couples capacitor 12 between the battery voltage and output node26. The combination of the closed states of switches 50 and 54 similarlycouples capacitor 14 between the battery voltage and output node 26(i.e., in parallel with capacitor 12). Thus, in the second phaseconfiguration of the 1/2A mode, the capacitor circuit includingcapacitors 12 and 14 in parallel with each other charges with respect tooutput node 26.

The 1/2B mode variant of the second mode configuration is shown in FIGS.6A-B. The second mode configuration includes both the 1/2A and 1/2Bmodes, or sub-modes, to minimize the number of switches that changestate during switching from one mode to another, as described below.Although these sub-modes are included in the illustrative embodiment, inother embodiments such sub-modes need not be included.

As shown in FIG. 6A, in the first phase configuration of the 1/2B mode,the switches 42, 46, 48, 52 and 56 are open, and the switches 40, 44, 50and 54 are closed. The combination of the closed states of switches 40and 44 couples capacitor 12 between the battery voltage and output node26. The combination of the closed states of switches 50 and 54 similarlycouples capacitor 14 between the battery voltage and output node 26(i.e., in parallel with capacitor 12). Thus, in the second phaseconfiguration of the 1/2B mode, the capacitor circuit includingcapacitors 12 and 14 in parallel charge with respect to output node 26.

As shown in FIG. 6B, in the second phase configuration of the 1/2B mode,the switches 40, 44, 48, 50 and 54 are open, and the switches 42, 46, 52and 56 are closed. The combination of the closed states of switches 42and 46 couples capacitor 12 between ground and output node 26. Thecombination of the closed states of switches 52 and 56 similarly couplescapacitor 14 between ground and output node 26 (i.e., in parallel withcapacitor 12). Thus, in the second phase configuration of the 1/2B mode,the capacitor circuit including capacitors 12 and 14 in parallel witheach other discharges with respect to output node 26.

As illustrated in FIGS. 7A-B, in a third configuration, the switchmatrix 16 can interconnect capacitors 12 and 14 in either the firstphase configuration shown in FIG. 5A or the second phase configurationshown in FIG. 5B. This third configuration can be referred to herein asthe “2/3 mode” because operation in this mode is intended to result inan output voltage signal at output node 26 having a voltage level thatis nominally about two-thirds of the battery voltage V_BATT, accordingto some implementations.

As shown in FIG. 7A, in the first phase configuration of the 2/3 mode,the switches 42, 46, 48, 52 and 56 are open, and the switches 40, 44, 50and 54 are closed. The combination of the closed states of switches 40and 44 couples capacitor 12 between the battery voltage and output node26. The combination of the closed states of switches 50 and 54 similarlycouples capacitor 14 between the battery voltage and output node 26(i.e., in parallel with capacitor 12). Thus, in the first phaseconfiguration of the 2/3 mode, the capacitor circuit includingcapacitors 12 and 14 in parallel with each other charges with respect tooutput node 26.

As shown in FIG. 7B, in the second phase configuration of the 2/3 mode,the switches 40, 44, 46, 50, 52 and 54 are open, and the switches 42, 48and 56 are closed. The combination of the closed states of switches 42,48 and 56 couples capacitors 12 and 14 in series between ground andoutput node 26. Thus, in the second phase configuration of the 2/3 mode,the capacitor circuit including capacitors 12 and 14 in series with eachother discharges with respect to output node 26.

As illustrated in FIG. 8A, the comparator circuit 18 can include fourcomparators 58, 60, 62 and 64 and a voltage level generator includingfour resistors 66, 68, 70 and 72. The resistors 66, 68, 70, and 72 canbe connected in series with each other between a supply voltage, such asthe battery voltage, and ground. The values of resistors 66, 68, 70, and72 can be selected such that the voltage at a node 74 at a first inputof comparator 60 (e.g., the inverting input) is approximately 2/3 of thesupply voltage V_BATT, the voltage at a node 76 at a first input ofcomparator 62 is approximately 1/2 of supply voltage V_BATT, and thevoltage at a node 78 at a first input of comparator 64 is approximately1/3 of supply voltage V_BATT. The second input (e.g., the non-invertinginput) of each of comparators 60, 62 and 64 can be coupled to thereference voltage signal V_REF. In other implementations, the voltagelevel generator can generate any suitable voltage levels, for example,via a resistive voltage divider.

The reference voltage signal V_REF can be an analog baseband signal. Insome instances, the reference voltage signal V_REF can track an outputpower of a power amplifier, such as the power amplifiers 5 describedearlier. The reference voltage signal V_REF can be independent of apower supply signal, such as the battery voltage V_BATT. The referencevoltage signal V_REF can be a clean signal with relatively little noise.For instance, in some implementations, noise features can distort thereference voltage signal V_REF by no more than about 0.01% to 0.5% ofthe magnitude of the reference voltage signal V_REF. In someimplementations, noise features can distort the reference voltage signalV_REF by no more than about 0.05% of the magnitude of the referencevoltage signal V_REF, no more than about 0.1% of the magnitude of thereference voltage signal V_REF, no more than about 0.25% of themagnitude of the reference voltage signal V_REF, or no more than about0.5% of the magnitude of the reference voltage signal V_REF. In someembodiments, a bypass capacitor (for example, as described earlier) canreduce noise and/or maintain a low level of noise on the referencevoltage signal V_REF. Using the reference voltage signal V_REF can avoidsituations in which the mode can get trapped due to jitter associatedwith a signal provided to a comparator. The output voltage V_OUT cancharge with the battery voltage V_BATT while the reference voltagesignal V_REF is independent of the battery voltage V_BATT. Moreover,basing a comparison of the comparator circuit 18 on the referencevoltage signal V_REF can increase the speed at which the output voltageV_OUT is charged up. In various implementations, the reference voltagesignal V_REF can be filtered prior to being provided to the input of anycombination of the comparators 58, 60, 62, and 64.

The output V_23 of the comparator 60 being high can indicate that thereference voltage V_REF exceeds (e.g., is greater in magnitude than) 2/3V_BATT; the output V_12 of the comparator 62 being high can indicatethat the reference voltage V_REF exceeds 1/2 V_BATT; and the output V_13of the comparator 64 being high can indicate that the reference voltageV_REF exceeds 1/3 V_BATT. One input of comparator 58 (e.g., theinverting input) can be connected to the output voltage signal V_OUT atthe output node 26. However, the other input of the comparator 58 (e.g.,the non-inverting input) can be connected to the reference voltagesignal V_REF. Thus, the output V_UD (or V_(UD)) of the comparator 58being high can indicate that the reference voltage V_REF exceeds theoutput voltage V_OUT. Conversely, the output V_UD of comparator 58 beinglow can indicate that the output voltage V_OUT exceeds the referencevoltage V_REF. In some implementations, the output V_UD of thecomparator 58 can serve as a direction comparison signal, indicating toswitching logic 20 (FIGS. 3A and/or 3B) in which direction, “up” or“down,” switching logic 20 should cause the output voltage signal V_OUTto change.

The outputs of one or more of the comparators 58, 60, 62, 64 can beprovided to a switch matrix, such as the switch matrix 16 describedearlier. In some implementations, the outputs of the comparators may beprovided to the switch matrix 16 via one or more intervening circuitelements.

Another example switch matrix in combination with capacitive circuitelements is shown in FIG. 8B. The switch matrix 16 illustrated in FIG.8B can implement any combination of features of the switch matricesdescribed herein. As shown in FIG. 8B, switches in a switch matrix 16can be represented by two or more sub-switches in parallel and/or inseries with each other. Each sub-switch can include a voltage controlledswitch, for example, an NFET and/or a PFET device. The switch matrix 16can be implemented with more than two capacitive circuit elements, suchas capacitors. For instance, as illustrated in FIG. 8B, the switchmatrix 16 can be implemented with four capacitors C1, C2, C3, and C4.The capacitors C1, C2, C3, and C4 can have approximately the samecapacitance in some implementations. According to other implementations,two or more of the capacitors C1, C2, C3, and C4 can have differentcapacitances. One or more of the capacitors C1, C2, C3, and C4 can havea different capacitance than the capacitor 28.

In some implementations, one or more switches in the switch matrix canbe implemented by a plurality of sub-switches in parallel. Each of theplurality of sub-switches can be controlled by a different input signal.For example, sub-switch sw1_1 can be controlled by output V_23 of thecomparator 60, sub-switch sw1_2 can be controlled by the output V_12 ofthe comparator 62, and sub-switch sw1_3 can be controlled by the outputV_13 of the comparator 64. As another example, sub-switch sw3_1 andsub-switch sw3_2 can be controlled by two different signals selectedfrom V_23, V_12, and V_13. The switch sw4 can be considered to have onlyone sub-switch, which can be controlled by V_13, V_12, or V_23. As alsoillustrated in FIG. 8B, some switches can include different numbers ofsub-switches (e.g., one, two, or three sub-switches as illustrated)based on design considerations. While four comparators are shown in FIG.8A, any suitable number of comparators can be included in order toachieve a desired level of granularity to control the switch matrix 16.For example, more comparators can be added to the comparator circuit 18to generate additional control signals to achieve finer resolution forswitch control over the range of voltage levels of the reference voltagesignal V_REF. In some of these implementations, more than threesub-switches that are controlled by different control signals generatedby the comparator circuit 18 may be provided for one or more switches ofthe switch matrix 16.

With the voltage level of the reference voltage signal V_REF changing,the output voltages of the comparators (e.g., V_23, V_12, and V_13) canbe used to dynamically adjust the effective switch size of switches inthe switch matrix 16 to improve and/or optimize efficiency at differentvoltage levels of the reference voltage signal V_REF. For example, at arelatively high voltage level (e.g., above 2/3 Vbatt, assuming a 1 to 1ratio between V_REF and V_OUT) of the reference voltage signal V_REF,all sub-switches in parallel can be turned on to lower the resistance ofthe switches in an on state. As another example, at a relatively lowvoltage level (e.g., below 1/3 Vbatt, assuming a 1 to 1 ratio betweenV_REF and V_OUT) of the reference voltage signal V_REF, only a portionof the sub-switches in parallel with each other can be turned on toreduce dynamic switching current. Thus, an effective size of a switch,such as a field effect transistor, in the switch matrix 16 can beadjusted based on current needed. As a result, less current can beconsumed dynamically charging one or more capacitors, such as capacitors12 and/or 14.

Switching frequency can also be controlled based on one or more outputsof the comparators 58, 60, 62, and 64. The effective resistance of theswitched capacitor network can be proportional to the reciprocal ofswitching frequency multiplied by capacitance. More sub-switches inparallel can reduce the effective resistance and fewer sub-switches inparallel can increase effective resistance of the switched capacitornetwork. Since dynamic power can be proportional to capacitancemultiplied by voltage squared multiplied by frequency, adjusting theswitching frequency can reduce power and/or enable selection of aswitching frequency for efficiency with a particular voltage level ofthe reference voltage signal V_REF. This can achieve better efficiencyfor specific voltage levels of the reference voltage signal V_REF.

FIG. 8C illustrates an example schematic of a bias control circuit for apower amplifier 5. The DC-DC voltage converter 214 of FIG. 8C caninclude any combination of features of the voltage converters describedherein, such as the DC-DC voltage converter 10. One or more of theoutputs from the comparator circuit 18, such as the comparators 60, 62,64 of FIG. 8A, can be used to dynamically adjust bias current for apower amplifier 5 based on a voltage level of the reference voltagesignal V_REF. Any suitable number of comparators can be included in thecomparator circuit 18 of the control logic in order to achieve a desiredlevel of resolution to control bias current of the power amplifier. Forinstance, more comparators can be included in the comparator circuit 18to achieve finer resolution for the power amplifier bias control overthe range of voltage levels of the reference voltage signal V_REF. Thiscan further improve the power amplifier efficiency with adjacent channelpower ratio (ACPR) margin. In some implementations, the bias currentprofile of the power amplifier 5 can track the voltage reference signaland/or the power amplifier power output level continuously, as well asthe collector voltage of the power amplifier 5.

In some implementations, the mode selection logic 32 of the switchinglogic 20 (FIGS. 3A and/or 3B) can include combinational logic configuredto determine the mode to which the switching logic 20 causes the switchmatrix 16 to switch in order to cause the output voltage signal V_OUT tochange in the direction indicated by the direction comparison signal,which can be generated by, for example, the comparator 58. The modeselection logic 32 can receive the comparison signals 30, which can bethe outputs of the comparator circuit 18 (for example, outputs of thecomparators 58-64) and/or based on the outputs of comparator circuit 18(for example, based on outputs of the comparators 58-64). The comparisonsignals 30 can be provided as inputs to the combinational logic. Thecombinational logic can be provided in any suitable form, such as anetwork of logic gates. For illustrative purposes, the combinationallogic will be described herein with reference to the table 80 shown inFIG. 9A. It will be understood that there are a number of ways toimplement the logic functions represented by the table 80 with a networkof logic gates or any other suitable form. The mode selection logic 32can output mode selection signals 36 (FIGS. 3A and/or 3B) in response tothe comparison signals 30 and the combinational logic.

As illustrated in FIG. 9A, Table 80 indicates the “next mode” (which canalso be referred to as a “next state”) to which switching logic 20 cancause the switch matrix 16 to switch in response to a combination of theoutputs V_UD, V_23, V_12 and V_13 of comparators 58, 60, 62, and 64,respectively. The modes indicated in table 80 are described above: the1/3 mode, the 1/2A mode, the 1/2B mode, and the 2/3 mode. Table 80 alsoindicates whether to “hold” the current mode, i.e., to maintain thecurrent mode as the next mode. Specifically, the outputs of all ofcomparators 58-64 being low can indicate that the current mode is to beheld in the (second phase configuration of the) 1/3 mode. In otherinstances, table 80 indicates that the mode is to switch. As describedbelow, the mode can switch from the current mode to the next mode onevery other clock cycle. It should be noted that a reference herein to“switching” or “changing” modes or to providing a mode control signal isintended to encompass within its scope of meaning not only changing to adifferent mode but also to maintaining the same mode at the time duringwhich mode switching can occur, i.e., switching or changing from thecurrent mode to the “next” mode in an instance in which both the currentmode and next mode are the same.

The mode selection logic 32 (FIGS. 3A and/or 3B) can include encodinglogic to encode some or all of the output, e.g., the next mode, andprovide mode selection signals 36 in an encoded form. The encoding logiccan encode the output in the form of, for example, a 3-bit word(MODE[2:0]). For example, the next mode output “1/3” can be encoded as“001”; the next mode output “1/2A” can be encoded as “010”; the nextmode output “1/2B” can be encoded as “011”; and the next mode output“2/3” can be encoded as “100”. As providing such encoding logic is wellwithin the capabilities of persons skilled in the art, it is not shownor described in further detail herein.

FIG. 9B is an example signal graph 900 illustrating the application ofthe example mode selection logic reflected in the truth table of FIG.9A. The line 902 represents the output voltage, V_OUT, of the DC-DCvoltage converter in the line 904 represents the reference voltage,V_REF, supplied to the DC-DC voltage converter. Further, the line 906represents the voltage supplied by the voltage supply, V_BATT. The lines908, 910, and 912 illustrate the points on the graph 900 correspondingto 1/3, 1/2, and 2/3 of the supply voltage, respectively. In the exampleillustrated in FIG. 9B, the reference voltage is a step function thatbegins slightly below 2/3 V_BATT and drops down to slightly above 1/3V_BATT after a period of time. It should be noted that although theexample reference voltage is illustrated as a step function, thereference voltage is not limited as such and a variety of types offunctions may be used to represent the reference voltage.

As illustrated by the line 914, the signal V_UD corresponding to thecomparison between the output voltage of the converter and the referencevoltage may have a ‘1’ value or logically high-value when V_OUT, or theoutput voltage, is less than V_REF or the reference voltage. When theoutput voltage exceeds the reference voltage, the V_UD value drops to‘0’ or a logic low value. It should be understood that whether the V_UDvalue is a ‘0’ or a ‘1’ when the output voltage is greater than thereference voltage is a matter of convention and is not limited to thevalues specified in the described examples herein. Similarly, the valuesof the additional comparators described herein are also not limited tothe example values presented.

In some implementations, the V_UD signal changes value only at aparticular point within a clock signal, such as the leading or risingedge of a clock signal. However, in other implementations, asillustrated by V_UD line 914 and the clock line 916 representing theclock signal, the V_UD value may change instantaneously upon a change inthe relationship between the output voltage and the reference voltageregardless of the clock signal.

As discussed above, the next mode to which the switching logic 20 cancause the switch matrix 16 to switch can occur in response to acombination of the outputs V_UD, V_23, V_12 and V_13 of comparators 58,60, 62, and 64, respectively. Further, as stated above, the line 914reflects the value for the output V_UD, which correlates to whether theoutput voltage V_OUT exceeds the desired voltage, or the referencevoltage. Similarly, the lines 918, 920, and 922 reflect the outputs ofV_13, V_12, and V_23, respectively. As seen from the graph 900, when theoutput voltage exceeds 1/3 V_BATT, the line 918 moves from logically lowto logically high. Similarly, when the output voltage exceeds 1/2V_BATT, the line 920 moves from logically low to logically high.Further, when the reference voltage is reduced to less than 1/2 V_BATTcausing the output voltage to fall below 1/2 V_BATT, the output of thecomparator V_12 moves from logically high back to logically low. Theline 922 remains logically low in the example illustrated in FIG. 9B asthe reference voltage does not reach 2/3 V_BATT and thus the outputvoltage does not reach 2/3 V_BATT.

As illustrated in FIG. 10, the switch control logic 34 can receive themode selection signals 36, which may be in the above-described encodedform of a 3-bit word (MODE[2:0]) and the “hold” signal. The MODE[2:0]word and “hold” signal together can indicate the next mode to whichswitching logic 20 is to switch. The “hold” signal can be latched into astate element, such as a flip-flop 82, in the control logic 34. TheMODE[2] bit can be latched into a state element, such as a flip-flop 84,in the control logic 34. The MODE[1] bit can be latched into a stateelement, such as a flip-flop 86, in the control logic 34. The MODE[0]bit can be latched into a state element, such as a flip-flop 88, in thecontrol logic 34. The flip-flops 82-88 can latch their inputs, on everyother cycle of the clock signal (CLOCK). Another flip-flop 90 can dividethe clock signal by two and provide the divided clock signal to theclock inputs of flip-flops 82-88. Other state elements can be usedinstead of flip-flops, such as latches and the like.

The switch control logic 34 also includes decoder logic 92 coupled tothe outputs of flip-flops 82-88. The decoder logic 92 can decode thelatched MODE[2:0] word and “hold” signal into the individual switchcontrol signals 38 (for example, S1-S9) that control the above-describedswitches 40-56 of the switch matrix 16 and/or any of the sub-switches inthe switch matrix 16. While mode selection signals 36 can indicate the“next” mode, the latched MODE[2:0] word and “hold” signal can indicatethe “current” mode. The decoder logic 92 can generate switch controlsignals 38 (for example, S1-S9) in response to the current mode and theclock signal.

The operation of some implementations of the decoder logic 92 isreflected in the circuit diagrams of FIGS. 4A-7B. It will be understoodthat for each mode configuration illustrated in FIGS. 4A-7B, theswitches 40-56 assume the first phase configuration during one half ofeach clock cycle and assume the second phase configuration during theother half of each clock cycle. In response to the latched MODE[2:0]word indicating the 1/3 mode or “001,” the decoder logic 92 can generateswitch control signals 38 (for example, S1-S9) to set the switches 40-56to the states shown in FIG. 4A during the first half of each clock cycleand to the states shown in FIG. 4B during the second half of each clockcycle. In response to the latched MODE[2:0] word indicating the 1/2Amode or “010,” the decoder logic 92 can generate switch control signals38 (for example, S1-S9) to set switches 40-56 to the states shown inFIG. 5A during the first half of each clock cycle and to the statesshown in FIG. 5B during the second half of each clock cycle. In responseto the latched MODE[2:0] word indicating the 1/2B mode or “011,” thedecoder logic 92 can generate switch control signals 38 (for example,S1-S9) to set switches 40-56 to the states shown in FIG. 6A during thefirst half of each clock cycle and to the states shown in FIG. 6B duringthe second half of each clock cycle. In response to the latchedMODE[2:0] word indicating the 2/3 mode or “100,” the decoder logic 92can generate switch control signals 38 (for example, S1-S9) to setswitches 40-56 to the states shown in FIG. 7A during the first half ofeach clock cycle and to the states shown in FIG. 7B during the secondhalf of each clock cycle. In response to the latched “hold” signalindicating the “hold” mode, the decoder logic 92 can generate switchcontrol signals 38 (S1-S9) to maintain switches 40-56 in their previousmode configurations during each half of the next clock cycle. It will beunderstood that any combination of the states described above can beimplemented in the opposite halves of each clock cycle and/or fordifferent portions/multiples of a clock cycle.

FIGS. 11 and 12 illustrate example methods of toggling between differentmodes of operation in the DC-DC converters 214 described herein.Efficiency can be improved by stopping the clock in response to V_OUTcrossing V_REF, thereby halting the switching network 16. While DC-DCconverters corresponding to the graph shown in FIG. 11 and/or the flowdiagram of FIG. 12 may include a specific number of modes forillustrative purposes, it will be understood that the principles andadvantages described with reference to FIGS. 11 and/or 12 can be appliedto a system with any suitable number of modes. Similarly, it will beunderstood that the threshold voltage levels (e.g., 1/3 Vcc, 1/2 Vcc,2/3 Vcc) are also described for illustrative purposes, and any suitablevoltage levels can be implemented with any combination of featuresdescribed in reference to FIGS. 11 and/or 12.

FIG. 11 is a timing diagram illustrating an example operation of theDC-DC voltage converter 214 of FIGS. 3A and/or 3B. The timing diagramshows the reference voltage signal V_REF and the output voltage signalV_OUT in one embodiment. In FIG. 11, the voltage level of the outputvoltage signal V_OUT over time is represented by the dashed curve andthe voltage level of the reference voltage signal V_REF over time isrepresented by the solid curve. As illustrated, the reference voltagesignal V_REF is a relatively noise free signal. The output voltagesignal V_OUT can track the reference voltage signal V_REF as thereference voltage signal V_REF changes via a DC-DC switched capacitorvoltage converter. The Vcc voltage is provided by the battery in thegraph shown in FIG. 11.

Depending on whether the output voltage signal V_OUT or the referencevoltage signal V_REF has a higher voltage level, the DC-DC converter mayoperate in a pulse skipping mode (PSM). For example, when the referencevoltage V_REF crosses 1/3 Vcc, the 1/3 mode may not be capable ofincreasing the output voltage V_OUT to be approximately equal to thereference voltage V_REF. As a result, the DC-DC converter 214 can switchto 1/2 mode. However, since the reference voltage V_REF may be justslightly higher than 1/3 Vcc, when changing to 1/2 mode, V_OUT mayovershoot the reference voltage V_REF and draw substantial current, suchas 3.0 mA, during an overshoot time period. In some cases, the overshootvoltage may depend on the load on the circuit. In some cases, theovershoot may be up to 2× the reference voltage. This can degrade theefficiency of the DC-DC converter 214. However, some of the embodimentspresented herein can reduce or prevent the occurrence of the overshootvoltage and the consequential substantial current draw that may beassociated with overshooting the reference voltage.

To address the problem of overshoot voltage, among others, the DC-DCconverter 214 can operate in pulse skipping mode (PSM) and/or adjusteffective switch size. In PSM mode, once the output voltage V_OUT passesthe reference voltage V_REF (for example, V_OUT becomes greater thanV_REF), the DC-DC converter 214 can turn-off an oscillator, let a loaddrain the current until the output voltage V_OUT becomes less than thereference voltage V_REF, and restart the clock to charge the outputvoltage V_OUT. Adjusting the switch size can be implemented, forexample, as shown and described with reference to FIG. 8B. For example,based on the load current when the output voltage V_OUT just passes thereference voltage V_REF, at least a portion of the sub-switches can beturned off to increase an effective resistance of one or more of theswitches. This can also help offset the current overshoot drawing from asupply voltage, such as Vbatt. This can also further increase the powerefficiency.

Referring back to FIG. 11, in some implementations, when the outputvoltage V_OUT is greater than the reference voltage V_REF, the DC-DCconverter 214 can operate in PSM mode. More specifically, while thereference voltage V_REF is below 1/3 Vcc, the mode control logic cankeep the DC-DC converter 214 in the 1/3 mode when the output voltageV_OUT is less than the reference voltage V_REF. When the output voltageV_OUT is greater than the reference voltage V_REF, the DC-DC converter214 can operate in the PSM mode instead of the 1/3 mode. When thereference voltage V_REF and the output voltage V_OUT are substantiallyequal, then the DC-DC converter 214 may operate in either mode,depending on the implementation. However, compare circuitry, such as thecomparator circuitry 18, can determine small differences between theoutput voltage V_OUT and the reference voltage V_REF.

When the voltage level of the reference voltage signal V_REF exceeds 1/3Vcc, the reference voltage V_OUT may not be able to exceed the referencevoltage V_REF in 1/3 mode. As a result, the control logic 20 can toggleone or more switches in the switch matrix 16 such that the DC-DCconverter 214 switches into the 1/2 mode. When V_REF is between 1/3 Vccand 1/2 Vcc, the DC-DC converter 214 can remain in 1/2 mode when V_OUTis less than V_REF and transition to the PSM mode when V_OUT is greaterthan V_REF. Similarly, when V_REF is between 1/2 Vcc and 2/3 Vcc, theDC-DC converter 214 can operate in 2/3 mode when V_OUT is less thanV_REF and operate in the PSM mode when V_OUT is greater than V_REF.Likewise, when V_REF is greater than 2/3 Vcc, the DC-DC converter 214can operate in supply mode or Vbatt mode when V_OUT is less than V_REFand operate in the PSM mode when V_OUT is greater than V_REF. Each timethe voltage level of the output voltage V_OUT crosses the referencevoltage signal V_REF, the control logic 20 can cause the DC-DC converter214 to operate in the PSM mode. Modes can be adjusted by toggling one ormore switches in the switch matrix 16 such that the DC-DC converter 214adjusts the mode.

As also shown in FIG. 11, ramp down can be limited by discharging aholding capacitor. In some implementations, it can be desirable for thecollector voltage of a power amplifier 5 that uses the DC-DC converter214 for biasing to be more than about 0.7 V. In addition, gapsseparating the reference voltage signal V_REF and the output voltagesignal V_OUT can be reduced in some instances, by increasing switch sizeso as to reduce IR drops at increasing currents.

FIG. 12 is a flow diagram illustrating an example process 1200 ofoperating the DC-DC voltage converter 214 of FIGS. 3A and/or 3B. Avoltage converter can operate in a mode, for example, any of the modesdescribed herein based on outputs generated from comparisons in acompare circuit, such as the compare circuit 18. For instance, one ormore comparisons performed by the comparators of FIG. 8A can be used todetermine a mode in which the DC-DC voltage converter 214 can operate.

In some implementations, the DC-DC voltage converter 214 can operate ina 1/3 mode when a reference voltage signal V_REF is less than 1/3 of asupply voltage. For instance, when the comparison at block 1202indicates that V_REF is less than 1/3 of the supply voltage and thecomparison at block 1204 indicates that the output voltage V_OUT is lessthan the reference voltage V_REF, the DC-DC converter 214 can be set to1/3 mode at block 1206.

The DC-DC voltage converter 214 can operate in a 1/2 mode when areference voltage signal V_REF is greater than 1/3 of a supply voltageand less than 1/2 of the supply voltage. For instance, when thecomparison at block 1210 indicates that V_REF is between 1/3 of thesupply voltage and 1/2 of the supply voltage and the comparison at block1212 indicates that the output voltage V_OUT is less than the referencevoltage V_REF, the DC-DC converter 214 can be set to 1/2 mode at block1214.

The DC-DC voltage converter 214 can operate in a 2/3 mode when areference voltage signal V_REF is greater than 1/2 of a supply voltageand less than 2/3 of the supply voltage. For instance, when thecomparison at block 1218 indicates that V_REF is between 1/2 of thesupply voltage and 2/3 of the supply voltage and the comparison at block1220 indicates that the output voltage V_OUT is less than the referencevoltage V_REF, then the DC-DC converter 214 can be set to 2/3 mode atblock 1222.

The DC-DC voltage converter can operate in a supply mode when areference voltage signal V_REF is greater than 2/3 of a supply voltage.For instance, when the comparison at block 1218 indicates that V_REF isbetween more than 2/3 of the supply voltage and the comparison at block1226 indicates that the output voltage V_OUT is less than the referencevoltage V_REF, then the DC-DC converter 214 can be set to Vbatt mode atblock 1228.

Based on one or more comparisons in the compare circuit 18, the DC-DCconverter 214 can also operate in PSM mode. For example, when the outputvoltage V_OUT is greater than the reference voltage V_REF, the DC-DCconverter 214 can operate in a PSM mode. For example, based on thecomparisons at blocks 1204, 1212, 1220, or 1226, the DC-DC converter 214can be set to pulse skipping mode at blocks 1208, 1216, 1224, or 1230,respectively.

In some implementations, the DC-DC converter 214 may switch to bypassmode in response to the reference voltage V_REF reaching a certainpercentage of Vcc. For example, as shown in FIG. 12, when the referencevoltage V_REF reaches 2/3 Vcc, the DC-DC converter 214 can operate inVbatt mode. The operations in FIG. 12 can be implemented concurrently,sequentially, or in any order, as appropriate.

PSM and FM Combination Overview

As previously described, in some embodiments, a DC-DC voltage convertercan be designed to operate in a pulse skipping mode (PSM). When V_OUTexceeds V-REF for a DC-DC switched capacitor with PSM capability, theDC-DC voltage converter enters PSM mode, or enables PSM, which mayresult in an oscillator (e.g., the oscillator 24) being turned off. Aspreviously described, in some such cases, a load may be used to drainthe current until the resulting V_OUT drops below V_REF. Further, theswitching network (e.g., switch matrix 16), used to modify V_OUT, mayalso be deactivated.

The Switched Capacitor DC-DC converter (e.g., the DC-DC voltageconverter 214) may be capable of reducing the battery voltage in fixedratios such as 1/2, 1/3, etc. This can be an efficient method ofgenerating the output voltage because there is a current gain equal tothe inverse of the voltage gain. There may be two phases for eachswitching mode (gain ratio)—charging and discharging the flyingcapacitors—and in normal operation, the clock duty cycle for each ofthese phases is just under 50%. However, there may be a short amount ofnon-overlap time when all switches to the flying capacitors are closed,resulting in duty cycles under 50%. Flying capacitors generally refersto capacitors that may float with respect to ground or a common node. Inother words, in some configurations of a circuit, the flying capacitorsmay not be electrically connected to or may be isolated from ground.These flying capacitors may instead be electrically connected betweentwo nodes other than ground. For example, with reference to FIG. 7A, thecapacitor 12 is electrically connected between V-BATT and V_OUT, but isisolated from the ground node illustrated in FIG. 7A. In other words, aflying capacitor may be shifted to connect between different nodes otherthan ground. A non-flying capacitor may be between a node, such asV_OUT, and ground.

In some cases, there is an associated output resistance with eachvoltage ratio that is a function of three modulation parameters:switching frequency, the resistances of the switches, and the duty cycleof each phase. The output resistance is inversely proportional tofrequency and directly proportional to the switch resistances andcomposite duty cycle. The output voltage follows the formula:V_OUT=V_BATT*Av−R_OUT*I_OUT, where Av is the voltage gain and I_OUT isthe current pulled by the load. Due to the output resistance, the outputvoltage cannot reach V_BATT*Av. For a continuous output design where thedesired output voltage can be set at a particular I_OUT, the outputresistance may be modulated by one of the three parameters mentionedabove.

In certain embodiments, once V_OUT falls below V_REF, PSM mode isdeactivated, or disabled, and the oscillator may be restarted orreinitialized. Further, the switching network may need to bereactivated. In some implementations, it may take a non-negligibleperiod of time to reactivate or restart the oscillator, associated clockgenerator, and related systems (e.g., switching network). For example,it may take up to 10 μs to begin operating the switch network afterreactivating the clock before V_OUT begins to catch up with V_REF. Insome communication systems, this 10 μs restart time can result in a callbeing dropped. Further, because the oscillator requires time to restart,there may be a delay in reactivating the switching network. Moreover,activating and deactivating the oscillator can create an effectiveswitching frequency that is higher than a manufacturer intendedfrequency of the oscillator. This higher frequency causes a reduction inthe power efficiency of the system.

In some implementations, a signal comparing V_OUT to the referencevoltage is used to turn off the oscillator when the output voltagebecomes too high. This allows the load to pull down the output voltage.In terms of the modulation parameters, the duty cycle of one phaserelative to the other phase would no longer be 1:1 but would rise tovalues as high as 15:1, and the frequency would rise by as much 140%before dropping to roughly 50% of the original frequency. When frequencyrises, the current used to drive the switches also rises, reducingefficiency.

In certain embodiments of the present disclosure, the DC-DC voltageconverter can be configured to use frequency modulation (FM) in additionto PSM. The FM may be implemented by using the amplified error signal todrive a voltage to a current conversion stage, and to drive a voltagecontrolled oscillator to dynamically adjust switching frequency based onthe V_REF level. Advantageously, in certain embodiments, by using FMwhen operating in PSM mode, the oscillator of the DC-DC voltageconverter may be kept operational eliminating the process ofreinitializing the oscillator when exiting PSM mode and improving theefficiency of the converter. In other words, in certain implementations,operation of the DC-DC voltage converter can be improved by reducing oreliminating the amount of time for initializing or reinitializing theoscillator before switching between voltage modes (e.g., switchingbetween 1/2 Vcc and 2/3 Vcc) and deactivating PSM. Further, in certainembodiments, using both PSM and FM can be particularly effective at lowpower regions where V_OUT and/or I_OUT are low because the drivercurrent can play a much larger role in the efficiency of the DC-DCvoltage converter.

Example Voltage Converter Using PSM and FM

FIG. 13 is a schematic block diagram of a voltage converter 1300, suchas a DC-DC voltage converter, that can operate using both PSM and FM inaccordance with certain embodiments. The voltage converter 1300 mayinclude some or all of the embodiments previously described with respectto the DC-DC voltage converter 214. For example, similar to theembodiments of the DC-DC voltage converter 214 illustrated in FIG. 2 andin FIGS. 3A and 3B, the voltage converter 1300 may include a switcharray or switch matrix 1302. This switch matrix 1302 may include anumber of switches that may be closed and opened so as to charge anddischarge a number of capacitive elements, or capacitors, included aspart of the switch matrix 1302. As with the switch matrix 16, the numberand type of capacitors that may be included by the switch matrix 1302 isnot limited. For example the switch matrix 1302 may include twocapacitors, three capacitors, ten capacitors etc. The output of theswitch matrix 1302, V_OUT, may be provided to a system or circuitry thatoperates using direct current at a particular voltage level. Forexample, the output voltage, V_OUT, of the switch matrix 1302 may beprovided to a power amplifier (e.g., the power amplifier die 204 or apower amplifier 5). In some embodiments, filter circuitry, such as acapacitor 1316, may be provided at the output node 1318. This filtercircuitry may be used to filter the output voltage signal V_OUT.

The voltage converter 1300 may further include control logic or switchlogic 1304 for controlling when the switches of the switch matrix 1302are opened or closed. As previously described with respect to thecontrol logic 17, the switch logic 1304 may include a comparator circuitsimilar to the comparator circuit 18 for comparing a reference voltage,V_REF, with an output voltage, V_OUT, output by the switch matrix 1302.The control signals of the switching logic 1304 may be based at leastpartially on the comparison between the V_REF and the V_OUT.

Further, the control signals output by the switch logic 1304 may beprovided to a driver 1306 before being provided to the switch matrix1302. The driver 1306 may be an inverter driver that is composed of aseries of inverter circuits and is designed to have an optimized drivingability stage by stage. Although the driver 1306, in some cases, may beformed from inverters, the driver 1306 is typically not intended toinvert the signal, but to boost the driving capability. However, in somecases, the driver 1306 may be designed to invert the signal. Theinverter circuits may be formed from pFET and nFETs. The driver 1306 canprovide the capability of driving the large capacitive loads of theswitch gates in the switch matrix 1302. Typically, the purpose of thedriver 1306 is to boost driving capability, rather than to invertsignals.

The voltage converter 1300 may further include a clock signal generator1308 which may generate a number of clock signals (e.g., clock signals1312A and 1312B) based on a signal provided or generated by anoscillator 1310. In certain embodiments, the clock signal generator 1308and the oscillator 1310 may include some or all of the embodimentsdescribed with respect to the clock signal generator 22 and associatedoscillator 24 in FIG. 3A or FIG. 3B. However, in contrast to theoscillator 24 that is activated by the enable signal, the oscillator1310 receives a signal from a voltage controlled current source (VCCS).Although not limited as such, the VCCS may be an operationaltrans-conductance amplifier (OTA) 1314, as illustrated in FIG. 13. TheOTA 1314 may cause a frequency generated by the oscillator 1310 to bemodulated based on a current provided by the OTA 1314. The currentprovided by the OTA 1314 may be based on a delta or a difference betweenthe V_REF and the V_OUT (V_REF−V_OUT). The greater the delta between theV_REF and the V_OUT, the greater the current produced by the OTA. WhenV_OUT>V_REF, delta V may be negative and the oscillator typically runsat its minimum frequency. Thus, the magnitude of delta V may effectivelyhave little to no impact on the oscillator frequency beyond triggeringthe minimum frequency in this case. However, when V_REF>V_OUT, delta Vmay be positive and can be used to set the OTA current, which can set orcorrespond to the oscillator frequency.

During operation, the OTA 1314 may cause the oscillator 1310 to reduceits generated frequency when V_OUT exceeds V_REF. On the other hand,when V_REF exceeds V_OUT, the OTA 1314 may cause the oscillator 1310 toincrease its generated frequency. In other words, in certain cases,frequency modulation may be performed based on the relationship betweenV_REF and V_OUT.

In general, the current produced by the OTA 1314 may be proportional tothe difference between V_OUT and V_REF, or ΔV. As the frequency may bedirectly proportional to the current input, the oscillator frequency canbe considered proportional to ΔV, or delta V. The oscillator frequencymay be kept within a continuous range. Further, the current, and thusthe oscillator frequency may have a maximum and a non-zero positiveminimum. When V_OUT>V_REF, the OTA current may be set to its minimumvalue, and therefore the oscillator 1310 may be set to its minimumnon-zero operating frequency. If the frequency is at a minimum but V_OUTremains greater than V_REF, PSM may be used in parallel with the FM tofurther reduce V_OUT. Further, a higher frequency can reduce outputimpedance so that V_OUT more closely approaches or tracks V_REF underhigh output current conditions. A lower frequency can improve powerefficiency by reducing the charging and discharging of the switchcapacitances, which can be relatively large resulting in a sizeablewasted amount of current. Thus, advantageously, reducing output currentas V_OUT approaches V_REF or when V_OUT exceeds V_REF can save power.

As previously described, the DC-DC voltage converter may include anumber of comparators that may be used to determine the switchconfigurations for the switches of the switch matrix. One of thesecomparators, for example the comparator 1320, may compare the V_OUT withthe V_REF. The output of this comparison may be referred to as V_UD.When the output voltage, V_OUT, of the DC-DC voltage converter 1300 isless than the reference voltage, V_REF, the V_UD has a logical onevalue. In some such cases, the switch matrix 1302 is configured suchthat the capacitors of the switch matrix 1302 continue to charge.Further, when the V_UD has a logical one value, the switch matrix 1302is configured such that the output voltage, V_OUT, continues to rise.The charging of the capacitors and the increasing output voltage may bedue to a source voltage supply, V_BATT (not shown), supplied to theDC-DC voltage converter 1300.

When the V_UD value is high, or logically one, the DC-DC voltageconverter disables PSM and the frequency of the oscillator signal ismodulated (e.g., FM is performed) by the OTA based on the differencebetween the V_OUT and V_REF values. The amount that the frequency ismodulated, although typically related to delta V, may beapplication-specific or environmental-specific. On the other hand, whenthe V_UD value is low, or logically zero, the DC-DC voltage converterenables PSM and the oscillator may be configured to run at a minimumoperational frequency, which may be application-specific orenvironment-specific. In some cases when V_UD is logic zero, theswitches of the switch matrix may be configured to allow the capacitorsof the switch matrix to drain and the supply voltage to be electricallydisconnected from the switch matrix. Further, as the supply voltage iscut off and the capacitors drain, the output voltage is reduced. Inaddition, the oscillator signal may be reduced to a lower frequency bythe OTA 1314. Once, the output voltage falls below the referencevoltage, V_UD may become logically one and the frequency of theoscillator signal may be restored to a higher operational frequency.Further, PSM may be disabled.

As stated above, the frequency modulation may be related to the amountof difference between the output voltage and the reference voltage. Forexample, in cases when V_OUT<V_REF, as V_OUT increases, the differencebetween V_REF and V_OUT decreases (e.g., V_REF−V_OUT decreases). As thedifference between the reference voltage and output voltage decreases,the output current of the OTA decreases. Consequently, the frequency ofthe oscillator decreases. This reduction in frequency in relation to thereduction of the output current from the OTA can serve as a form of FMcontrol via the voltage controlled current source (VCCS), which isimplemented by the OTA.

As the output voltage increases and approaches the reference voltage,the oscillator frequency decreases. Advantageously, in certainembodiments, the reduction in the oscillator frequency results inimproved power efficiency for the DC-DC converter because as thedifference between the reference voltage and the output voltagedecreases, the DC-DC voltage converter uses less energy to furtherincrease the output voltage. Further, once the output voltage surpassesthe reference voltage (V_OUT>V_REF), the oscillator can be set to run ata low or minimum operating frequency. This minimum operating frequencycan be a minimum signal frequency that enables the oscillator tocontinue operating instead of being turned off. In some cases, theminimum operating frequency may be based on the oscillator or may beapplication-specific. Thus, in some cases, when V_OUT<V_REF, the DC-DCconverter operates in a FM mode that uses a continuously adjusted ormodified frequency. Further, when V_OUT>V_REF, the DC-DC converter mayoperate in a PSM mode with the oscillator operating at a minimumoperating frequency.

In some implementations, one or more of the comparators included in thecontrol logic of the DC-DC voltage converter may be hysteresiscomparators. For example, the comparator 1320 may be a hysteresiscomparator. Advantageously, in certain embodiments, using a hysteresiscomparator reduces or prevents continuous or rapid mode changes of theswitch matrix 1302 or the DC-DC voltage converter 1300. The rapid modechanges may occur because of noise or because the output voltage hoversaround the reference voltage. For example, the DC-DC voltage converter1300 may be configured to operate in one mode when the output voltageexceeds the reference voltage and may be configured to operate in asecond mode when the output voltage falls below the reference voltage.As the output voltage oscillates between a point above the referencevoltage and a point below the reference voltage, there may be continuousor rapid mode changes of the DC-DC voltage converter 1300 and in somecases, the DC-DC voltage converter 1300 may not complete entering afirst mode before being reconfigured to enter a second mode. Thus, thehysteresis comparator 1320 enables more stable transitions between modesof the DC-DC voltage converter 1300 and prevents undesirable rapidchanges between modes of the DC-DC voltage converter 1300. In certainembodiments, the hysteresis comparator 1302 may enable more stabletransitions by, for example, introducing a small hysteresis voltage.This small hysteresis voltage can be used to prevent rapid fluctuationsby, for example, requiring a voltage that surpasses a threshold, whichmay cause a state change, to drop below the threshold by at least thehysteresis voltage level before reverting to the previous state ortransitioning to another state.

Typically, the hysteresis value applied to the hysteresis comparator1320 is set to a few millivolts. In other cases, the hysteresis valuemay relate to a number of clock cycles that a voltage should be below athreshold before triggering a state change. However, the hysteresisvalue of the hysteresis comparator 1320 may be selected based on aminimum period of time that the DC-DC voltage converter 1300 shouldremain in a particular state. Some cases, minimum period of time may beone clock cycle or a number of clock cycles required to initialize acomponent of the DC-DC voltage converter 1300, such as the oscillator1310 or one or more switches of the switch matrix 1302.

As illustrated in FIG. 13, in some embodiments the oscillator 1310 mayinclude logic. This logic can be used to determine whether theoscillator 1310 is to generate a signal with a frequency based on acurrent received from the OTA 1314 or whether the oscillator 1310 shouldgenerate some other frequency signal, such as a signal with a minimumoperating frequency for the DC-DC voltage converter 1300. Althoughillustrated as one system, in some implementations the logic may beseparate from the oscillator 1310. A number of implementations for thelogic are possible. For example, the oscillator 1310 may include amultiplexor that can output a minimum frequency signal or a signal witha frequency based on a current from the OTA 1314 based on the V_UDsignal, or some other control signal. As a second example, theoscillator 1310 may include one or more combinational gates forcontrolling whether an output of the oscillator is a minimal operatingfrequency or a frequency based on the output of the OTA 1314. In someembodiments, the output of the OTA 1314 may serve as the control signalfor the oscillator 1310.

In some implementations, when the V_UD signal is high, indicating thatthe output voltage of the DC-DC voltage converter 1300 is below thereference voltage, the output signal from the oscillator 1310 may have afrequency determined based on a current output by the OTA 1314. Thus, aform of frequency modulation may occur where the frequency of the outputsignal is directly proportional to the current output by the OTA 1314.Further, the current output by the OTA 1314 may be based on thedifference between the V_REF and V_OUT signals. In some cases, thegreater the difference between the V_REF and V_OUT signals, the higherthe frequency of the signal output by the oscillator 1310. In someembodiments, the oscillator 1310 may have a maximum frequency value. Insome such cases, once the difference between the V_REF and V_OUT signalsreaches a threshold, the oscillator 1310 may be set to output a signalat the maximum frequency.

However, when V_UD is low and PSM is enabled, the output of theoscillator 1310 may be set to a minimum operating frequency regardlessof the current provided by the OTA 1314. In some such cases, the systemmay be designed to cause the oscillator to run at a minimum operatingfrequency for the purpose of saving power while maintaining theoscillator 1310 in an active state. In such embodiments, when V_UD ishigh after a period of being low, the oscillator does not need to bereactivated because the oscillator 1310 remained in an active state.Thus, in certain embodiments, transitions between modes of the DC-DCvoltage converter 1300 may be performed more efficiently and faster.

Second Example Voltage Converter Using PSM and FM

FIG. 14A is another schematic block diagram of a voltage converter thatcan operate using both PSM and FM according to certain embodiments. Thevoltage converter 1400 provides similar functionality as the DC-DCvoltage converter 1300, but is illustrated using a design similar tothat of the DC-DC voltage converter 214. Thus, elements of the voltageconverter 1400 that are in common with the DC-DC voltage converter 1300and/or the DC-DC voltage converter 214 share the same referencenumerals.

The voltage converter 1400 includes control logic 17 that can controlthe switches of the switch matrix 16. Further, the control logic 17 canprovide a control signal (e.g., V_UD or V_(UD)) to the FM controller1410. The FM controller 1410 can use the control signal in combinationwith an output voltage signal to determine a frequency modulated (FM)oscillator signal, which can be provided to the clock generator 22.Advantageously, the FM controller 1410 can modify the oscillator signaland consequently the clock signal generated by the clock generator 22within a desired frequency range. Although in some cases the FMcontroller 1410 can deactivate the oscillator and provide no signal, ora constant value, typically the FM controller 1410 varies the FMoscillator signal between a minimum frequency and a maximum frequency.Often, the minimum frequency is selected to be a minimum operatingfrequency that enables the oscillator to continue operating withoutbeing deactivated or without requiring the oscillator to be reinitiatedprior to exiting a PSM mode for the DC-DC voltage converter 1400.

FIG. 14B is a schematic block diagram illustrating additional details ofthe voltage converter presented in FIG. 14A. The voltage converter 1450is an embodiment of the voltage converter 1400 that illustratesadditional details of the voltage converter 1400. As with the voltageconverter 1400, the voltage converter 1450 provides similarfunctionality as the DC-DC voltage converter 1300, but is illustratedusing a design similar to that of the DC-DC voltage converter 214. Thus,elements of the voltage converter 1450 that are in common with the DC-DCvoltage converter 1300 and/or the DC-DC voltage converter 214 share thesame reference numerals.

As illustrated in FIG. 14B, in contrast to the enable signal that isprovided to the DC-DC voltage converter 214 of FIGS. 3A and 3B, afrequency modulated (FM) oscillator signal is provided to the clockgenerator 22 of the DC-DC voltage converter 1450. The FM oscillatorsignal may vary based at least in part on whether the output voltageexceeds the reference voltage. In some implementations, the V_CTRL1 andV_CTRL2 inputs may be omitted. For example, the V_CTRL1 and V_CTRL2inputs may be replaced by the outputs of the comparator circuit 18.These outputs can be based on the V_REF signal.

The FM controller 1410, which is represented by a dashed line box,includes the OTA 1314 and the oscillator 1310, which were bothpreviously described above. As previously described, the OTA 1314 canmodulate the frequency of the signal output by the oscillator 1310,which is provided to the clock generator 22. Further, the comparatorcircuit 18 can provide a control signal to the oscillator 1310 that canbe used to determine whether the oscillator 1310 is set to a minimumoperating frequency.

In some embodiments, the DC-DC voltage converter 1450 may receive anenable signal, or alternatively, a disable signal. This enable signalmay be received over an enable input 1412 or enable pin, which may beincluded by the FM controller 1410. Alternatively, the enable input 1412may be a disable input. The signal received over the enable input 1412may be used to shut down the DC-DC voltage converter 1450. Asillustrated in FIG. 14B, the signal from the enable input 1412 may beprovided to various elements of the DC-DC voltage converter 1450including the OTA 1314, the oscillator 1310, and the DC-DC converterblock 214 and used to shut down or deactivate the various elements.Alternatively, the lack of receiving the signal may be used to shut downor deactivate the various elements of the DC-DC voltage converter 1450.

Driver Circuit Current

FIG. 15 is a graph 1500 comparing the difference in driver circuitcurrent drawn by a voltage converter operating with PSM and a voltageconverter operating with PSM and FM, such as the DC-DC voltage converter1450. Typically, the switches of the switch matrix 16 are very large andtherefore have a large gate capacitance on the order of a few hundredpF. Turning the switches on and off may require charging and dischargingthese large capacitances by the driver circuit. The current may beproportional to switching frequency and capacitance, or switch size.Typically, the current being drawn is not going to the output load, butis instead dissipated as heat. Thus, the efficiency of the DC-DC voltageconverter is reduced. As illustrated by the graph 1500, a design of aDC-DC voltage converter that uses both PSM and FM has a reduced drivercurrent across a set of output voltages compared to a voltage converterthat utilizes only PSM. Thus, a design that uses both PSM and FM is moreefficient than a design that utilizes only PSM.

Further, as illustrated by the line 1502, the current of a DC-DC voltageconverter that utilizes only PSM spikes around 1 volt, 1.5 volts, and 2volts and can reach over 3.5 mA. As illustrated by the line 1504, aDC-DC voltage converter that utilizes both PSM and FM also has currentspikes around 1 volt, 1.5 volts, and 2 volts. However, as illustrated bythe graph 1500, the current spikes are less than 50% of the currentspikes of the DC-DC voltage converter that utilizes only PSM. Thus, theDC-DC voltage converter that utilizes both PSM and FM has a much greaterefficiency than the DC-DC voltage converter that does not implement FMboth across the range of voltage values and at voltage points associatedwith a high current. The voltage points with the high current values, orcurrent spikes, may be attributed to points of high switching frequencywithin the switch matrix relating to a high frequency of turning theoscillator on and off.

FIG. 16 is a graph 1600 comparing the power efficiency of a DC-DCvoltage converter operating with PSM with a DC-DC voltage converteroperating with PSM and FM. The line 1602 represents the efficiency of aDC-DC voltage converter operating with PSM. The line 1604 represents theefficiency of a DC-DC voltage converter operating with PSM and FM. Ascan be seen by comparing line 1602 with line 1604, the efficiency isroughly the same or better across a range of reference voltages. Inother words, despite the oscillator continuing to be powered for theDC-DC voltage converter that operates with PSM and FM, the efficiency ofthe DC-DC voltage converter remains comparable to or better than theDC-DC voltage converter that deactivates its oscillator when not in use.In particular, at low power ranges (e.g., when V_REF<1.0V), theefficiency is improved with the PSM and FM implementation compared tothe PSM only implementation of the DC-DC converter.

Applications

Some of the embodiments described above have provided examples inconnection with wireless devices, such as mobile phones, that includepower amplifiers. However, the principles and advantages of theembodiments can be used for any other systems or apparatus that have aneed for DC-DC voltage conversion.

Such systems with DC-DC converters can be implemented in a variety ofelectronic devices including, for example, consumer electronic products,parts of the consumer electronic products, electronic test equipment,etc. The consumer electronic products can include, but are not limitedto, a mobile phone (e.g., a smart phone), a telephone, a television, acomputer monitor, a computer, a hand-held computer, a tablet computer, alaptop computer, a personal digital assistant (PDA), a microwave, arefrigerator, an automobile, a stereo system, a cassette recorder orplayer, a DVD player, a CD player, a VCR, an MP3 player, a radio, acamcorder, a camera, a digital camera, a portable memory chip, a washer,a dryer, a washer/dryer, a copier, a facsimile machine, a scanner, amulti-functional peripheral device, a wrist watch, a clock, etc. Partsof the consumer electronic products that can include a DC-DC converteras described herein can include a multi-chip module, a power amplifiermodule, an integrated circuit including a DC-DC converter, etc.Moreover, other examples of the electronic devices can also include, butare not limited to, memory chips, memory modules, circuits of opticalnetworks or other communication networks, and disk driver circuits.Further, the electronic devices can include unfinished products.

Additional Embodiments

One aspect of this disclosure is an apparatus that includes a switchmatrix and control logic. The switch matrix can include switchesconfigurable into a plurality of states corresponding to a plurality ofvoltage levels of an output voltage. The switch matrix may be configuredto adjust a state of at least one of the switches based at least in parton one or more mode control signals. The control logic may have a firstinput coupled to a first voltage generated from a voltage supply and asecond input coupled to a reference voltage independent of the voltagesupply. The control logic may be configured to compare the first voltagegenerated from the voltage supply with the reference voltage independentof the voltage supply. The control logic may be further configured togenerate the one or more mode control signals based at least in part onthe comparison.

The apparatus can also include a plurality of capacitive circuitelements operatively coupled to the switch matrix. The switch matrix canbe configured to adjust electrical connections between the capacitivecircuit elements and an output node. Two or more of the switches in theswitch matrix can be controlled by different mode control signals. Theswitch matrix can also be configured to implement a plurality of modes,each mode having a first phase configuration in which at least one ofthe plurality of capacitive circuit element is charged and a secondphase configuration in which the at least one of the plurality ofcapacitive circuit element is discharged.

The control logic can also be configured to compare the referencevoltage to a second reference voltage generated from the power supply,in which the second reference voltage generated from the power supplyhas a different potential difference than the first voltage generatedfrom the power supply. The one or more mode control signals can begenerated based on comparing the reference voltage to the secondreference voltage generated from the power supply. The control logic canalso be configured to compare the reference voltage to a third voltagegenerated from the power supply, in which the third reference voltagegenerated from the power supply has a different potential differencethan the first voltage generated from the power supply and the secondvoltage generated from the power supply. The one or more mode controlsignals can be generated based on comparing the reference voltage to thethird voltage generated from the power supply. The control logic can beprogrammable between continuous and discrete voltage modes. In somecases, the output voltage may not be filtered by an inductor of an LCfilter.

The reference voltage can track an output power of a power amplifier.The apparatus can also include a bypass capacitor having a first endcoupled to the reference voltage and a second end coupled to analogground. Noise on the second reference voltage can be at least about anorder of magnitude less than noise on the output voltage.

Another aspect of the disclosure is a multi-chip module that includes apower amplifier die including one or more power amplifiers and acontroller die. The controller die may include a power amplifier biascontrol and a direct current to direct current (DC-DC) converter. TheDC-DC converter may be configured to generate an output voltage based onone or more mode control signals. Further, the DC-DC converter may havemode control logic configured to compare a plurality of voltagesgenerated from a voltage supply to a reference voltage independent ofthe voltage supply and to generate the one or more mode control signalsbased at least in part on the comparisons.

The power amplifier die can include a GaAs device and the controller diecan include a CMOS device. The power amplifier can include a CDMA poweramplifier, such as a dual-band CDMA power amplifier or a tri-band CDMApower amplifier. The multi-chip module can be configured to be mountedon a phone board.

Another aspect of this disclosure is a mobile device that includes abattery, a power amplifier, and a direct current to direct current(DC-DC) voltage converter. The battery can be configured to power themobile device. The power amplifier may be configured to amplify a radiofrequency (RF) input signal and to generate an amplified RF outputsignal. The DC-DC voltage converter can be configured to generate anoutput voltage to control the power amplifier so as to improve powerefficiency. The DC-DC voltage converter can include a switch matrixhaving a plurality of mode configurations corresponding to a pluralityof output voltage levels. The switch matrix may be configured to adjusta state of one or more switches based at least in part on one or moremode control signals. The DC-DC voltage converter can also includecontrol logic configured to compare a first voltage generated from thebattery to a reference voltage indicative of an output power of thepower amplifier, and to generate the one or more mode control signalsbased at least in part on the comparison. The DC-DC voltage convertercan be configured to transition into pulse skipping mode based at leastin part on the comparison.

At least one of the one or more switches can have a plurality ofsub-switches, and at least one of the plurality of sub-switches can beconfigured to change a state based at least in part on an amount ofcurrent provided by the DC-DC voltage converter. Further, the mobiledevice can be configured to communicate using, for example, a 3Gcommunications standard or a 4G communications standard. The mobiledevice can be configured, for example, as a smart phone or a tabletcomputer.

Another aspect of this disclosure is an apparatus that includes a switchmatrix and control logic. The switch matrix can include switchesconfigurable into a plurality of states corresponding to a plurality ofvoltage levels of an output voltage. The switch matrix may be configuredto adjust a state of the plurality of states based at least in part on amode control signal. The control logic may have a first input coupled toa voltage generated from a voltage supply and a second input coupled toa low noise voltage having noise features causing a distortion with amagnitude of no more than about 0.1% of a magnitude of the low noisesignal voltage. The control logic may be configured to compare thevoltage generated from the voltage supply with the low noise voltage andto generate the one or more mode control signals based at least in parton the comparison.

The low noise voltage can have noise that is one or two orders ofmagnitude less than the output voltage. The low noise voltage can havenoise features causing a distortion with a magnitude of no more thanabout 0.05% of a magnitude of the low noise signal voltage

Yet another aspect of this disclosure is a method that includes:comparing a first voltage generated from a voltage supply with areference voltage independent of the voltage supply; generating a modecontrol signal based at least in part on the comparing; and adjusting astate of at least one switch in a switch matrix based on the modecontrol signal to adjust a voltage level of an output of the switchmatrix. The method can also include additionally comparing a voltageindicative of the output of the switch matrix with the referencevoltage, and generating the mode control signal can also be based on theadditionally comparing. Moreover, the method can also includeadditionally comparing the reference voltage independent of the voltagesupply with a second voltage generated from the voltage supply andhaving a different voltage level than the first voltage generated fromthe voltage supply, and generating the mode control signal can also bebased on the additionally comparing.

CONCLUSION

Unless the context clearly requires otherwise, throughout thedescription and the claims, the words “comprise,” “comprising,” and thelike are to be construed in an inclusive sense, as opposed to anexclusive or exhaustive sense; that is to say, in the sense of“including, but not limited to.” The word “coupled”, as generally usedherein, refers to two or more elements that may be either directlyconnected, or connected by way of one or more intermediate elements.Additionally, the words “herein,” “above,” “below,” and words of similarimport, when used in this application, shall refer to this applicationas a whole and not to any particular portions of this application. Wherethe context permits, words in the above Detailed Description using thesingular or plural number may also include the plural or singular numberrespectively. The word “or” in reference to a list of two or more items,that word covers all of the following interpretations of the word: anyof the items in the list, all of the items in the list, and anycombination of the items in the list.

Moreover, conditional language used herein, such as, among others,“can,” “could,” “might,” “e.g.,” “for example,” “such as” and the like,unless specifically stated otherwise, or otherwise understood within thecontext as used, is generally intended to convey that certainembodiments include, while other embodiments do not include, certainfeatures, elements and/or states. Thus, such conditional language is notgenerally intended to imply that features, elements and/or states are inany way required for one or more embodiments or that one or moreembodiments necessarily include logic for deciding, with or withoutauthor input or prompting, whether these features, elements and/orstates are included or are to be performed in any particular embodiment.

A number of signals are described herein as having logic low values andlogic high values. Further, the functions of a number of systems aredescribed as depending on the logic value of particular signals. Itshould be understood that in some cases the logic values selected arefor example and that it is possible for the systems to function asdescribed using reverse logic values. For instance, in some cases, V_UDmay be set to logic high when V_OUT exceeds V_REF and V_UD may be set tologic low when V_REF exceeds V_OUT. In such cases, control circuitry maybe modified to process the alternative logic signals.

The above detailed description of embodiments is not intended to beexhaustive or to limit the invention to the precise form disclosedabove. While specific embodiments of, and examples for, the inventionare described above for illustrative purposes, various equivalentmodifications are possible within the scope of the invention, as thoseskilled in the relevant art will recognize. For example, while processesor blocks are presented in a given order, alternative embodiments mayperform routines having acts, or employ systems having blocks, in adifferent order, and some processes or blocks may be deleted, moved,added, subdivided, combined, and/or modified. Each of these processes orblocks may be implemented in a variety of different ways. Also, whileprocesses or blocks are at times shown as being performed in series,these processes or blocks may instead be performed in parallel, or maybe performed at different times.

The teachings provided herein can be applied to other systems, notnecessarily the system described above. The elements and acts of thevarious embodiments described above can be combined to provide furtherembodiments.

While certain embodiments of the inventions have been described, theseembodiments have been presented by way of example only, and are notintended to limit the scope of the disclosure. Indeed, the novel methodsand systems described herein may be embodied in a variety of otherforms; furthermore, various omissions, substitutions and changes in theform of the methods and systems described herein may be made withoutdeparting from the spirit of the disclosure. The accompanying claims andtheir equivalents are intended to cover such forms or modifications aswould fall within the scope and spirit of the disclosure.

What is claimed is:
 1. A voltage converter comprising: a switch matrixincluding a set of switches configurable into a plurality of statescorresponding to a plurality of voltage levels, the switch matrixconfigured to adjust a state of at least one of the set of switchesbased at least in part on one or more switch control signals and tooutput a first voltage; an oscillator configured to generate anoscillator signal at a frequency; a clock generator configured togenerate a clock signal based on the oscillator signal; and controllogic configured to produce the one or more switch control signals basedat least in part on a comparison between the first voltage and a secondvoltage that is supplied as an input to the voltage converter, thecontrol logic further configured to, based at least in part on thecomparison between the first voltage and the second voltage, generate afrequency control signal for modifying the frequency of the oscillator.2. The voltage converter of claim 1 further comprising a frequencymodulation (FM) controller configured to modify the frequency of theoscillator to generate a modified oscillator signal.
 3. The voltageconverter of claim 2 wherein the FM controller is further configured tosupply the modified oscillator signal to the clock generator.
 4. Thevoltage converter of claim 2 wherein the FM controller is furtherconfigured to modify the frequency of the oscillator based at least inpart on the comparison between the first voltage and the second voltage.5. The voltage converter of claim 4 wherein the FM controller is furtherconfigured to reduce the frequency of the oscillator as the differencebetween the first voltage and the second voltage decreases.
 6. Thevoltage converter of claim 4 wherein the FM controller is furtherconfigured to reduce the frequency to a default frequency upon the firstvoltage exceeding the second voltage.
 7. The voltage converter of claim6 wherein the default frequency corresponds to a threshold frequency foroperation of the oscillator.
 8. The voltage converter of claim 2 whereinthe FM controller includes a voltage controlled current source (VCCS)configured to provide a current signal to the oscillator.
 9. The voltageconverter of claim 8 wherein the current signal is based at least inpart on the comparison between the first voltage and the second voltage.10. The voltage converter of claim 1 further comprising at least onecapacitor in electrical communication with the switch matrix, the atleast one capacitor configured to enable the plurality of voltagelevels.
 11. The voltage converter of claim 10 wherein the capacitor ischarged or discharged based at least in part on a configured state fromthe plurality of states of the switch matrix.
 12. A wireless devicecomprising: a battery configured to power the wireless device; a poweramplifier configured to amplify a radio frequency (RF) input signal andto generate an amplified RF output signal; and a voltage converterincluding a switch matrix, an oscillator, a clock generator, and controllogic, the switch including a set of switches configurable into aplurality of states corresponding to a plurality of voltage levels, theswitch matrix configured to adjust a state of at least one of the set ofswitches based at least in part on one or more switch control signalsand to output a first voltage, the oscillator configured to generate anoscillator signal at a frequency, the clock generator configured togenerate a clock signal based on the oscillator signal, and the controllogic configured to produce the one or more switch control signals basedat least in part on a comparison between the first voltage and a secondvoltage that is supplied as an input to the voltage converter, thecontrol logic further configured to, based at least in part on thecomparison between the first voltage and the second voltage, generate afrequency control signal for modifying the frequency of the oscillator.13. The wireless device of claim 12 wherein the set of switches areFETs.
 14. The wireless device of claim 12 wherein the voltage converterfurther includes a frequency modulation (FM) controller configured tomodify the frequency of the oscillator to generate a modified oscillatorsignal and to supply the modified signal to the clock generator.
 15. Thewireless device of claim 14 wherein the FM controller includes a voltagecontrolled current source (VCCS) configured to provide a current signalto the oscillator, the modified signal supplied to the clock generatoroutput by the oscillator and based at least in part on the currentsignal.
 16. The wireless device of claim 12 wherein the voltageconverter is configured to transition into a pulse skipping mode basedat least in part on the comparison of the first voltage with the secondvoltage.
 17. A multi-chip module comprising: a power amplifier dieincluding one or more power amplifiers; and a controller die including apower amplifier bias controller and a voltage converter, the voltageconverter including a switch matrix, an oscillator, a clock generator,and control logic, the switch including a set of switches configurableinto a plurality of states corresponding to a plurality of voltagelevels, the switch matrix configured to adjust a state of at least oneof the set of switches based at least in part on one or more switchcontrol signals and to output a first voltage, the oscillator configuredto generate an oscillator signal at a frequency, the clock generatorconfigured to generate a clock signal based on the oscillator signal,and the control logic configured to produce the one or more switchcontrol signals based at least in part on a comparison between the firstvoltage and a second voltage that is supplied as an input to the voltageconverter, the control logic further configured to, based at least inpart on the comparison between the first voltage and the second voltage,generate a frequency control signal for modifying the frequency of theoscillator.
 18. The multi-chip module of claim 17 wherein the voltageconverter further includes a frequency modulation (FM) controllerconfigured to modify the frequency of the oscillator to generate amodified oscillator signal and to supply the modified signal to theclock generator.
 19. The multi-chip module of claim 18 wherein the FMcontroller includes a voltage controlled current source (VCCS)configured to provide a current signal to the oscillator, the modifiedsignal supplied to the clock generator output by the oscillator andbased at least in part on the current signal.
 20. The multi-chip moduleof claim 19 wherein the voltage converter is configured to transitioninto a pulse skipping mode based at least in part on the comparison ofthe first voltage with the second voltage.